Fullscreen HTML5Med Res (??MB)HTML4

Teach-In 2026 by Mike Tooley World of Wireless – An Introduction to Radio and Wireless Technology, part 12.8 Frequency conversion with multipliers and mixers L ast month, we explored some useful techniques for analysing RF networks and devices, including S-parameters, vector network analysis (VNA), and spectrum analysis (SA). VNA plays a vital role in RF engineering by offering precise measurements of network and device properties across a wide frequency spectrum. Spectrum analysis is essential for examining the components within RF signals to evaluate spectral purity and identify spurious emissions. This month, our focus shifts to frequency conversion methods used in receivers, transmitters and synthesisers, with particular emphasis on how multipliers and mixers can change the frequency of a signal. A comprehensive understanding of circuit operation can be achieved Multiplier fin fin fin fin (×n) Multiplier Multiplier (×n) (×n) Multiplier (×n) through experiments conducted with software simulation. Accordingly, this month we will be introducing the use of SPICE software to investigate multiplier and mixer circuits. Our Hands-On project is a 50MHz Digital Frequency Meter that’s ideal for making a variety of RF measurements. Frequency conversion In radio and wireless, the ability to change or ‘convert’ the frequency of a signal is essential to the operation of all but the most basic types of receivers and transmitters. Frequency changers, such as multipliers and mixers, are the essential devices that make this possible. By shifting signals to higher or lower frequencies, they allow for easier filtering, improved selectivity, and enhanced system performance. fout = nfin fout = nfin fout = nfin fRF 4 fRF Mixer Mixer C1 1nF TR1 2N2222 Mixer (b) Mixers (b) Mixers Fig.8.1: the difference between a multiplier and a mixer. (b) Mixers (b) Mixers c b e fIF = (fLO±fRF) plus other fIF = (fLO±fRF) mixing products fIF = (fLO±fRFplus ) other mixing products plus other fIF =products (fLO±fRF) mixing plus other mixing products RF output (27MHz) C4 33pF R2 68kΩ RF input (9MHz) Mixer fRF T1 C3 100n LO fLO fRF +9V fout = nfin (a) Multipliers (a) Multipliers (a) Multipliers fLO (a) Multipliers f fLO Frequency conversion is fundamental to superheterodyne receivers (see Part 2, December 2025), where incoming signals are converted to a fixed intermediate frequency (IF). This permits more effective filtering, processing and demodulation. In transmitters, generated signals can be moved to frequencies suitable for transmission in a desired band. Understanding the principles and applications of frequency changers is therefore vital for anyone involved in RF design and wireless communications. Fig.8.1 shows the basic principles of the two types of frequency changer. In Fig.8.1(a) the input frequency, fin, is multiplied by a factor of n to produce an output frequency, fout, where fout = n × fin. Typical values of n range from two to five. For example, an arrangement R1 22kΩ R3 100Ω C2 10nF 0V Fig.8.2: the circuit of a simple frequency multiplier. Practical Electronics | June | 2026 Fig.8.3: a Tina-TI simulation of the circuit in Fig.8.2. where n = 2 (known as a ‘doubler’) will produce a signal at 50MHz when supplied with an input at 25MHz. In a similar manner, a circuit where n = 3 (known as a ‘tripler’) will produce a 21MHz signal when supplied with an input signal at 7MHz. In Fig.8.1, the intermediate frequency output. fIF, is generated by mixing the input frequency, fRF, with the local oscillator (LO) frequency, fLO. The mixing process is not simply additive as it is with a basic audio mixer, but instead relies on non-linear characteristics within the mixer. In additive mixing, two signals are combined by simply summing their voltages. The resulting output will contain both original frequencies, but no new frequency components are generated. By contrast, multiplicative mixing uses a non-linear device (such as a diode or transistor) to produce additional frequency components, as well as the sum and difference of the original frequencies. These multiplicative ‘mixer products’ generate signals at new frequencies, thereby allowing signals to be shifted to different frequency bands. In Fig.8.1(b), the intermediate frequency at the output of the mixer can be calculated as fIF = (fLO ± fRF). So, for example, if an IF of 455kHz is to be obtained from an RF input at 14MHz, the local oscillator would need to be running at either 14.455MHz (above the RF input) or 13.545MHz (below the RF input). Practical Electronics | June | 2026 Either of these two local oscillator frequencies will produce the required intermediate frequency, but there can be reasons to prefer one or the other. Popular intermediate frequencies are 455kHz and 470kHz (AM), or 1.6MHz, 9MHz, 10.7MHz and 21.4MHz (FM); the value chosen depends on the application, modulation method, and receiver complexity. It’s worth noting that some receivers use double and even tripleconversion techniques, in which case several mixers will be present. Multipliers A basic form of frequency multiplier is shown in Fig.8.2. This circuit uses a conventional NPN transistor connected in common-emitter mode. The collector load is a tuned circuit formed by the primary winding of T1 and capacitor C4. This paralleltuned circuit is resonant at 27MHz (three times the RF input frequency). To provide a better match and improve the Q-factor of the load, the collector of TR1 is connected to a tap in T1’s primary at about its midpoint. When a 9MHz input is applied to the base of TR1, the output voltage from the secondary of T1 will be reasonably sinusoidal at three times the input frequency (the circuit thus acts as a ‘tripler’). Fig.8.3 shows the Tina-TI simulation of Fig.8.2. For simplicity, we’ve replaced the tuned transformer with a simple LC tuned circuit (see Part 2 of Teach-In). The values of C1 and L1 (20pF and 1µH, respectively) are chosen so that the tuned circuit is resonant at approximately 27MHz. In Fig.8.3, we’ve adjusted the signal source, VG1, so that it produces a sinewave at 9MHz with an amplitude of 500mV. The resulting rectangular pulses of current at the collector of TR1 excite harmonic resonance in the collector tuned circuit, so the collector voltage will be a sinewave at three times the input frequency. In Fig.8.3, we’ve used Tina’s excellent virtual oscilloscope to display the input and output waveforms. Note that the input (shown in red) has a peak-to-peak value of about 1V, whereas the output (shown in green) is almost 20V peak-to-peak. Notice that some of the input frequency signal is present in the output waveform, but at a significantly lower level. The residual amplitude can be further reduced by increasing the Q-factor of the load, as in the transformer-coupled arrangement shown earlier in Fig.8.2. If you would like to experiment with the component values in Fig.8.3, it can be instructive to calculate values for L1 and C1 to make the circuit act first as a doubler (n = 2) and then as a quadrupler (n = 4). You can calculate the required values using the formula shown on page 42 in Teach-In Part 2: introducing RF components and circuits. Simply click on the components (L and C) in the simulation to change 5 +9V their values and observe the effect on the output waveform. If you don’t already have this excellent simulation tool, Tina-TI can be freely downloaded from https://www. ti.com/tool/TINA-TI R2 68kΩ Mixers Fig.8.4 shows a basic mixer. The RF input is applied to the base of TR1 via capacitor C1, while the local oscillator is coupled into the emitter by capacitor C2. The intermediate frequency output is extracted from the collector of TR1; C4 and the primary of T1 are resonant at the intermediate frequency (in this case, 450kHz). To remove the unwanted mixing products (eg, those resulting from signal components just above and below 3.5MHz) a filter is connected at the output of T1. This passive component accepts the intermediate frequency but rejects the other mixing products (of which there are many). IF filters can be based on LC networks, quartz crystals or low-cost ceramic resonators. Typical filter bandwidths are 15kHz for AM broadcasts, 150kHz for FM broadcasts or 9kHz for AM voice communication. Narrow-band modes, such as singlesideband (SSB), continuous wave (CW), and low-speed data (such as PSK31) all benefit from narrower filters with bandwidths of 3kHz or less. Fig.8.5 shows the Tina-TI simulation of Fig.8.4. For simplicity, we’ve once again replaced the tuned T1 C3 100n C4 390pF TR1 2N2222 C1 1nF IF filter (450kHz) c b RF input (3.5MHz) e R1 22kΩ IF output (450kHz) C2 10nF Local oscillator (3.95MHz) R3 470Ω 0V Fig.8.4: a basic mixer circuit diagram. transformer with a simple LC tuned circuit. The values of C3 and L1 (390pF and 330µH, respectively) are chosen so that the tuned circuit is resonant at approximately 450kHz. We’ve once again used Tina’s virtual oscilloscope to display the input and output waveforms. The RF signal source, VG1, should be adjusted so that it produces a sinewave at 3.5MHz with an amplitude of 500mV. To produce a 500kHz IF, the local oscillator source (VG2) needs to be set to 4MHz with an amplitude of 500mV. Note how the signal input (shown in red) has a peak-to-peak value of about 1V, whereas the output (shown in green) is once again around 20V peak-to-peak (indicating that there is some gain present in the mixer). It’s also worth noting that some of the local oscillator signal is present in the output waveform, but at a much lower level. The residual LO signal content can be further reduced by increasing the Q-factor of the load, as in the more selective transformer-coupled arrangement shown earlier in Fig.8.4. You might like to experiment with different RF signal input and local oscillator frequencies in the settings Fig.8.5: a Tina-TI simulation of the circuit in Fig.8.4. 6 Practical Electronics | June | 2026 +12V T1 C3 10n C2 100pF Local oscillator (15.6MHz) RF input (14MHz) R2 68kΩ C5 180pF C1 100pF R1 47kΩ g2 d g1 s R3 100kΩ IF output (1.6MHz) IF filter (1.6MHz) R4 220Ω TR1 BF961 C4 100nF 0V Fig.8.6: an improved mixer based on a BF961 dual-gate Mosfet. of the two voltage generators, VG1 and VG2. For example, it can be instructive to see if you can produce the same intermediate frequency with the local oscillator injection on the low side of the RF input (ie, set VG2 to 3MHz rather than 4MHz). You could also try a higher intermediate frequency of, say, 1.6MHz, with the local oscillator (VG2) set to 5.1MHz. Fig.8.6 shows the circuit of an improved mixer arrangement based on a dual-gate Mosfet. TR1 is connected in common-source mode with the RF signal input applied to gate 1 (G1) and the local oscillator applied to gate 2 (G2). Since both gates have a very high impedance, the circuit requires very little signal and local oscillator drive power. The values shown in Fig.8.5 are for operation at 14MHz with an IF at 1.6MHz. High-side local oscillator injection is provided at 15.6MHz. A practical implementation of Fig.8.6 is shown in Fig.8.7. With RF input and local oscillator frequencies of 7.2MHz and 7.655MHz respectively, this arrangement uses a low-cost ceramic filter to produce an IF at 455kHz. The LT455EU filter provides a 6dB bandwidth of ±7.5kHz and is thus ideal for AM broadcast reception. Additional filters in the LT455 series offer bandwidths as low as ±2kHz, making them a better choice for SSB reception. Image channel rejection As previously mentioned, a Local oscillator input mixer produces the same IF when it receives signals at two distinct RF input frequencies: one above and one below the local oscillator frequency. Of these, one represents the desired signal, while the other is unwanted and referred to as the ‘image channel’. The frequency difference between the desired signal and the image channel is twice the intermediate frequency. Image channel rejection in a superhet receiver is the ability of a receiver to suppress or reject signals that appear at the image frequency. Image channel rejection is usually specified in dB and is the ratio of the amplitudes of the wanted and image channel signals that produce the same IF output amplitude. Effective image channel rejection ensures that unwanted signals do not interfere with the reception of wanted signals, improving image channel selectivity and overall receiver performance. To mitigate the impact of signals present in the image channel, one or more stages of input filtering are essential. Increasing the intermediate frequency reduces the filtering requirements for the RF input stages. Consequently, a receiver can achieve significantly higher image channel rejection with an IF of 1.6MHz than with, for example, 455kHz. Typical values of image rejection range from around 40dB for an inexpensive AM superhet receiver to better than 70dB for a high-end communications receiver featuring dual conversion and two intermediate frequencies. 455kHz IF filter RF input DC supply Tuning voltage IF output BF961 dual-gate MOSFET mixer Practical Electronics | June | 2026 Fig.8.7: a practical implementation of a dual-gate Mosfet mixer, including a low-cost ceramic IF filter. 7 Local oscillator (59MHz) Diode mixers D1 BAT42 IF output (9MHz) T1 RF input (50MHz) T2 D2 BAT42 0V Fig.8.8: a simple diode balanced mixer. Local oscillator (59MHz) D1-D4 BAT42 T1 RF input (50MHz) T2 IF output (9MHz) 0V Fig.8.9: a double-balanced diode mixer. It is often easier to obtain the best image rejection when the LO is above the received frequency rather than below it because that shifts the image frequency above the broadcast band, in the case of an AM receiver, allowing a low-pass filter to reject it without affecting desired signals. If you look back at the Tina simulation of the simple mixer shown in Fig.8.5, you will see that an IF output of around 20V peak-to-peak is being produced by an input of 1V peak-topeak, corresponding to a conversion gain of 20 (or about 26dB). Amplification is more readily accomplished at lower frequencies, which is why the majority of gain in a superheterodyne receiver is provided within cascaded intermediate frequency (IF) stages (following the mixer). Gains typically range from 80100dB. As a result, mixers are generally not required to deliver substantial amounts of gain. Conversion gain Many mixers provide a small (but useful) amount of gain. This ‘conversion gain’ is simply the ratio of the IF output signal voltage amplitude to the RF input signal voltage. Conversion gain is usually expressed in dB, and typical values of conversion gain vary from about 12-26dB. Simple mixers can be based on low-cost switching diodes, as shown in Fig.8.8. Although a single diode can be used, the balanced arrangement reduces the level of local oscillator injection reaching the IF output. In the arrangement shown in Fig.8.8, the RF input is at 50MHz, with the local oscillator at 59MHz. The 9MHz output is selected by T2, which also provides impedance matching for the 9MHz filter at the output. Unlike the Mosfet mixer that we met earlier, this type of mixer requires significantly greater drive power. It also provides attenuation rather than the modest amount of conversion gain associated with the earlier mixers shown in Figs.8.4 and 8.6. Losses associated with diode mixers vary from about 6-9dB (ie, a ‘conversion loss’ rather than a ‘conversion gain’!). A double-balanced diode mixer is shown in Fig.8.9. This arrangement provides rejection of both the local oscillator and RF signal components present at the IF output. The two transformers are often untuned, and the broadband characteristic makes this type of mixer useful over a very wide range of frequencies. Monolithic mixers LSI technology is increasingly being used to create low-distortion, wide dynamic range mixers with excellent characteristics. One such example is the AD831, shown in Fig.8.10. Typical applications include RF-to-IF down-­ conversion in HF and VHF receivers, second mixers in digital mobile radio (DMR) base stations, directto-baseband conversion, quadrature modulation, demodulation and even Doppler shift detection in ultrasound imaging applications. Filter network Fig.8.10: a simplified internal schematic diagram of the AD831 monolithic mixer IC. AN IFN +V IFP 50Ω AP COM C2 RT RFP L1 100nF + - IF output OUT RFN 8 Set gain 110Ω 50Ω + RF input C1 51Ω VFB CC 100nF Mixer LO input LOP + - LON 51Ω 100nF Practical Electronics | June | 2026 The AD831 incorporates a low7.0-7.2MHz AF noise IF output amplifier as well as a local oscillator driver. It consists of a mixer core, a limiting ampli7.0-7.2MHz AF Bandpass AF fier, a low-noise output amplifier RF Mixer IF filter amplifier and a bias circuit. The device has differential RF and local oscillator LO Bandpass AF RF Mixer IF inputs: RFP, RFN and LOP, LON, filter amplifier RF 7.0-7.2MHz where the Ps and Ns indicate ‘phase’ LO and ‘antiphase’ respectively. RF The mixer’s differential outputs at 7.0-7.2MHz VFO the sum and difference frequencies of the RF and local oscillator inputs are available at IFP and IFN. These VFO open-collector outputs can supply (a) Direct conversion receiver for CW/SSB an external filter network (not shown in Fig.8.10) or can be connected to (a) Direct conversion receiver for CW/SSB internal loads that convert the dif- Fig.8.11: a ferential output currents into dif- low-cost mixer 7.0-7.2MHz 3.5-3.8MHz 455kHz ferential voltages that can be fed toAF module based on the AD831 chip. an external IF amplifier. 3.5-3.8MHz 455kHz When the RF 7.0-7.2MHz and LO inputs are AF Bandpass AF RF RF in IF AC-coupled (as shown Fig.8.10), RF IF IF out Mixer IF filter Mixer filter amplifier an on-chip network supplies bias oscillator, reducingamplifier problems 7.0-7.2MHz with moving the AF HF spectrum into the Bandpass AF LOoscillator current for the RF and local isolation and shielding, andRFimprov- LO VHF range RF Mixer IF RF IF for reception out IF filter using aIFlowMixer filter amplifier amplifier 7.0-7.2MHz AF (see Teach-In Part 4 stages. This network can be disabled ing overall system performance. As cost SDR dongle RF 7.0-7.2MHz 3.955-4.255kHz Bandpass LO to the when the mixer is DC-coupled previously mentioned, the RF, IF, RFonMixer software definedAFradio, February LO RF IF filter amplifier external RF and local oscillator stages. and LO ports may be DC-coupled 2026). The AD831 can also serve as a RF Bandpass AF RF LO IF 3.955-4.255kHz The internal low-noise amplifier7.0-7.2MHz and supplied fromRFa symmetrical second in cellular and two-way Mixermixer filter amplifier VFO VFO stage has inputs AP and AN. When ±5V supply or, more commonly, radio base stations, offering better perRF LO 7.0-7.2MHz the amplifier is used, the AN and AP AC-coupled and operated from a formance than conventional mixers. VFO VFO pins are supplied directly from pins single DC supply of 9-12V. RF IFN and IFP. on-chip 50Ωreceiver resis- for CW/SSB The AD831 operates with RF Applications 7.0-7.2MHz (a)Two Direct conversion (b) ConventionalVFO superhet receiver tors are then used as loads to convert and local oscillator inputs up to To aid in construction and experithe differential mixerconversion outputsreceiver into for 500MHz mentation, the AD831 is available (a) Direct CW/SSB and features an untuned (b) Conventional VFOsuperhet receiver differential input voltages for the differential output bandwidth of from several suppliers mounted internal low-noise IF amplifier. The up to 250MHz. This makes the chip on a small circuit board fitted with 3.5-3.8MHz 455kHz 14.0-14.35MHz 4.0-4.35MHz for CW/SSB (a) Direct conversion gain of this stage is set by external ideal for broadband applications SMA receiver connectors, as shown in feedback resistors connected to the is conversion Fig.8.11.receiver The board is configured for (a) Direct for CW/SSB 3.5-3.8MHz 455kHz where a large chunk of spectrum 14.0-14.35MHz 4.0-4.35MHz VFB pin.RF converted from oneBandpass broad frequency AC-coupled operation and requires a Tuneable IF RF MixertheIF need IF out IFrange filter to another. RF Mixer IF IFshows out Theamplifier driver eliminates DC supply of 9-12V. Fig.8.12 filter system 7.0-7.2MHz 3.5-3.8MHz for a high-power external local AF A typical applicationBandpass might be just a few 455kHz of the possibilities. RF Tuneable IF RF LO Mixer amplifier7.0-7.2MHz RF RF Bandpass filter Bandpass filter RF RF RF IF LO Mixer Mixer LO VFO IF IF LO VFO RF AF IF filter IF out RF 3.955-4.255kHz AF 7.0-7.2MHz 7.0-7.2MHz LO XOSC IF out VFO 4.0-4.35MHz (b) Conventional14.0-14.35MHz superhet receiver IF LO IF out RF RF Mixer IF out 3.955-4.255kHz 3.955-4.255kHz RF amplifier IF 14.0-14.35MHz RF Mixer LO 14.0-14.35MHz Bandpass filter Bandpass filter IF out (c) High-performance receiver with tuneable IF system VFO (c) High-performance receiver with tuneable IF system VFO RF VFO IF out 3.955-4.255kHz 3.955-4.255kHz Practical Electronics | June | 2026 IF out (b) Conventional14.0-14.35MHz superhet receiver 4.0-4.35MHz XOSC LO RF RF IF filter 10MHzIF filter IF filter 10MHz IF filter 14.0-14.35MHz 4.0-4.35MHz (b) Conventional superhet receiver Bandpass Tuneable IF RF Mixer IF filter system 3.5-3.8MHz 455kHz Tuneable IF Bandpass RF Mixer IF LO filter 3.5-3.8MHz system 455kHz 10MHz Mixer LO IF out system (c) High-performance receiver with tuneable IF system VFO (a)14.0-14.35MHz Direct conversion receiver for CW/SSB 4.0-4.35MHz LO 10MHz IF XOSC 14.0-14.35MHz 4.0-4.35MHz (a) Direct conversion receiver for CW/SSB Mixer IF RF Mixer IF RF XOSC 455kHz RF VFO RF LO Mixer IF (c) High-performance receiver with tuneable IF system VFO (b) Conventional VFO superhet receiver RF RF amplifier RF amplifier RF RF (b) Conventional superhet receiver RF 3.5-3.8MHz RF amplifier RF RF amplifier amplifier AF 3.955-4.255kHz amplifier Mixer RF LO filter LO RF IF 4.0-4.35MHz 4.0-4.35MHz 10MHz Mixer IF RF Mixer IF 10MHz XOSC LO LO XOSC RF out RF amplifier Tuneable IF system IF Tuneable system RF out IF out IF out Fig.8.12: some typical applications for the 10MHz AD831 mixer chip. 10MHz (d) Transmitter XOSC frequency generator (d) Transmitter frequency generator XOSC (c) High-performance receiver with tuneable IF system 9 In Fig.8.12(a), the AD831 is used in a simple direct-conversion receiver where the local oscillator operates at the same frequency as the incoming RF signal and modulation is converted directly to baseband audio frequency. In effect, this type of receiver has zero intermediate frequency! The bandpass filter improves receiver selectivity and reduces unwanted mixing of harmonics of the local oscillator with strong out-of-band signals. A conventional superhet receiver is shown in Fig.8.12(b). Here, the 455kHz intermediate frequency is obtained from the difference between the incoming RF signal and the variable-frequency local oscillator. The RF amplifier provides gain and selectivity, while the IF filter reduces the level of unwanted mixing products. Fig.8.12(c) shows an arrangement where the local oscillator is fixed and the IF amplifier is tuned over a range of frequencies. Circuits of this type are often found in dual-conversion receivers, where the first local oscillator is crystal-­ controlled (and therefore very stable), and a second, variable-frequency local oscillator provides tuning. Mixers are found in transmitters as well as receivers; Fig.8.12(d) shows a typical application where the output of a low-frequency VFO is mixed with that from a crystal oscillator to produce a variablefrequency output at a higher frequency. The AD831’s exceptional frequency response makes this arrangement ideal for multi-band frequency generation, where different crystals can be used for band selection. Fig.8.13: our 50MHz Digital Frequency Meter is built around this PIC-based kit. C1 10nF Input 74HC04 C2 10nF P1-2 1 D1 4.3V D2 4.3V IC1a 2 + C3 100nF 3, 5, 9, 11 R1 100kΩ 13 14 IC1f +6V P2-2 C4 47µF e 12 Output P2-3 7 R1 100kΩ GND P1-1 GND P2-1 Fig.8.14: our input protection and shaping circuit for the Digital Frequency Meter. Hands-On: A 50MHz Digital Frequency Meter This month’s Hands-On project is another useful item of RF test equipment, a portable 50MHz Digital Frequency Meter. The completed instrument is auto-ranging, uses a five-digit seven-segment LED display, and operates from an internal DC supply derived from four standard 1.5V alkaline cells. To reduce cost and simplify construction, we’ve based the project on an inexpensive PIC-based frequency counter (see Fig.8.13). The module is available in kit form from numerous online suppliers (just search for “DIY digital frequency counter crystal tester”). Construction and operation are straightforward, but to overcome 10 Figs.8.15 & 16: the component and track layouts for the input protection circuit. the lack of input protection (essential in a general-purpose test instrument), we’ve added a second circuit board to safeguard the in- put and provide some additional waveform shaping. The circuit of the input protection and shaping board is shown Practical Electronics | June | 2026 0V 1 14 2 13 3 12 4 11 5 10 6 9 7 8 +VCC Fig.8.17: the pin connections for IC1, a 74HC04 hex inverter. in Fig.8.14. Two zener diodes, D1 and D2, clamp the input voltage to a maximum peak-to-peak voltage of about 4.6V. The clamping arrangement is followed by a hex CMOS inverter stage with its input biased to half- supply voltage. A sinusoidal input signal of about 150mV peak-to-peak at 1-10MHz is sufficient to produce a logic-compatible signal at the output of IC1a with fast rise and fall times. The output of IC1a is further improved by a second inverting stage, IC1f. The unused inputs of the remaining four inverting stages are all held high and their outputs left disconnected. Assembly Instructions and assembly diagrams for the frequency counter and LED display board can be found on the internet at https:// www.zl2pd.com/xtalchecker.html The supplied PCB is clearly marked with component outlines and values, but care needs to be taken to ensure that the three transistors are identified and fitted correctly. The external power connector (marked “DC5-9V” on the PCB) should be fitted but is not used because the internal DC supply (+6V and GND from S1a and S1b respectively) is fed directly to the “+” and “-” pins on the four-way header. The mode button supplied Fig.8.19: wiring up the Frequency Meter. with the kit can also be fitted but is not required for normal operation of the Digital Frequency Meter. The component layout for the input protection and shaping circuit board is shown (viewed from the top) in Fig.8.15. The corresponding track layout (viewed from below) is illustrated in Fig.8.16. 29 track breaks are required; they can be made using a spot face cutter or small drill bit. Eleven links on the upper side of the boards can be made using short lengths of tinned copper wire. The pin connections for IC1 are shown in Fig.8.17. When the two circuit boards are complete, they should be mounted in the ABS enclosure using 10mm hex spacers and interconnected as shown in Figs.8.18 & 8.19. When positioning the two circuit boards, adequate clearance must be left below them to accommodate the battery holder in the base of the enclosure. Keep the signal and ground leads for the input connector (SK1) as short as possible. Testing and calibration The testing procedure is straightforward, but before inserting the cells into the holder and switching it on, it’s important to carry out a careful visual examination of both boards. Check the orientation of semiconductors such as diodes, transistors and integrated circuits. When the visual check has been S1a On/off SK1 Input + + P2-3 Input conditioning P2-2 P1-1 P2-1 board P1-2 IN GND B1 6v DFM board Fig.8.18: the interconnecting arrangement of the 50MHz Digital Frequency Meter. Practical Electronics | June | 2026 S1b 11 Sensitivity (mVRMS) 800 600 400 200 0 10k 100k 1M Frequency (Hz) 10M 100M Fig.8.21: the frequency response of our prototype 50MHz Digital Frequency Meter. completed, insert four 1.5V AA tivity fell below 50mV RMS over alkaline cells and switch the in- a frequency range extending from strument on. 600kHz to 10MHz, and was better With no signal applied, a single than 250mV RMS from 75kHz to zero should appear on the display 50MHz. (this reduces standby battery drain). When a signal is applied, IF offsetting the remaining segments will beThe low-cost frequency counter come illuminated and the supply has several features that can be current will increase. accessed using the pushbutton Since the instrument is ‘auto- switch, which can be made acranging’, the decimal point indica- cessible by drilling a small hole tion will change according to the in the enclosure before assembly. input frequency range. A momentary press of this button A small trimmer capacitor (ad- is needed to enter programming jacent to the 20MHz quartz crystal mode, after which you can toggle and 22pF capacitor) provides fre- through various options by further quency adjustment, but calibration presses. will not normally be necessary and To modify the settings, you need should not be attempted without the aid of an accurate Fig.8.22: the finished 50MHz Digital frequency reference. Frequency Meter. We will be describing an accurate 1MHz/10MHz frequency reference in the next instalment of Teach-In, so if you don’t have a reference source available, you might want to wait until next month before making any adjustment. The Digital Frequency Meter is ideal for making RF measurements; the measured response of our prototype instrument is shown in Fig.8.21. Our prototype meter produced useful indications up to around 75MHz. Its sensi12 to keep the button depressed for more than one second to apply the selection. The available options are “Quit”, “Add”, “Sub”, “Zero” and “Table”. Selecting these options will allow you to display the RF input frequency of a superhet receiver while the instrument measures the local oscillator frequency. The IF offset adjustment is then made automatically by subtraction or addition. This can be a very handy feature! To select an IF offset, you will first need to navigate to the “Table” option. Making a further button press for more than one second before releasing will allow you Practical Electronics | June | 2026 to cycle through the list of IF frequencies: 455.0kHz, 4.1943MHz, 4.4336MHz, 10.700MHz and 3.9990MHz. To select one of these, simply press the button for a further second before releasing it while the desired option is being displayed. The selected frequency can be either added to the reading (when the local oscillator is on the low side) or subtracted from it (when the local oscillator is on the high side) by returning to the main option list and selecting one of the “Add” or “Subtract” options. Coming up! In Part 9 next month, we take a deep dive into RF sources, explaining how to get the best out of a variety of signal and marker generator modules and direct digital synthesis (DDS) chips. Our Hands-On constructional project is an accurate calibration source and programmable frequency marker. Following that, part 10 will cover RF sources and oscillators, including temperature compensation, oven-controlled oscillators and direct digital synthesis (DDS). PE See you then! Parts List – 50MHz Frequency Meter 1 small ABS enclosure, 158 × 90 × 60mm or larger 1 25 × 64mm piece of stripboard (9 × 24 holes) 1 DIY frequency counter crystal tester kit (see text) 1 DPDT miniature toggle switch (S1) Connectors 1 2-pin male 0.1in/2.54mm header (P1) 1 3-pin male 0.1in/2.54mm header (P2) 1 14-pin low-profile DIL IC socket 1 chassis-mounting female BNC socket, mounting hardware and earthing tag (SK1) 1 battery holder for 4 AA cells 1 battery clip and connecting lead Hardware 8 brass or nylon M3 hexagonal 10mm tapped standoffs/spacers 4 M3 panhead machine screws 4 M3 brass hex nuts 8 M3 countersunk head machine screws Semiconductors 1 74HC04 CMOS hex inverter (IC1) 2 4.3V 300mW zener diodes (D1, D2) Capacitors 1 47µF 16V radial electrolytic (C4) 2 10nF 50V ceramic (C1, C2) 1 100nF 50V ceramic (C3) Resistors (all ¼W axial, ±5% or better) 2 100kΩ ¼W ±5% or better axial resistors (R1, R2) JTAG Connector Plugs Directly into PCB!! No Header! No Brainer! Our patented range of Plug-of-Nails™ spring-pin cables plug directly into a tiny footprint of pads and locating holes in your PCB, eliminating the need for a mating header. Save Cost & Space on Every PCB!! Solutions for: PIC . dsPIC . ARM . MSP430 . Atmel . Generic JTAG . Altera Xilinx . BDM . C2000 . SPY-BI-WIRE . SPI / IIC . Altium Mini-HDMI . & More www.PlugOfNails.com Tag-Connector footprints as small as 0.02 sq. inch (0.13 sq cm) Practical Electronics | June | 2026 13