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AUDIO OUT AUDIO OUT L R By Jake Rothman Discrete discoveries and a cascode amplifier PCB design Baxandall balanced input Since my cascode amplifier circuit can be configured as inverting, it seemed logical to try adding an input balancing transformer using the Baxandall configuration described last month. This led to some interesting effects. It has to be admitted this is a risky approach, since we are expecting the limited negative feedback to linearise three things at once: the output stage, output capacitor and input transformer. In analog design, it’s desirable to linearise one thing at a time with separate loops, hence the profusion of op amps used in modern designs. Trying to make negative feedback (NFB) do too much can lead to complex interactions and odd frequency effects. In the days of valves and expensive transistors, though, this approach was normal, with NFB being taken around multiple capacitors and transformers. To get this to work without the circuit bursting into low-frequency oscillation (‘motorboating’) involves a lot of component value tweaking. This is the sort of design challenge I enjoy, because you get more out of less. It’s more creative than stringing together standard circuits and is a privilege only afforded to the home constructor or sole trader today. Bumps in the frequency domain As expected, LF oscillation occurred at around 8Hz when adding the modem transformer to the input (as shown in Fig.4(a) in the August 2025 issue). This was soon tamed by changing the lowfrequency (LF) step network resistor, 70 R3, to 11kΩ. All that remained was a tolerable peak of 1.5dB at 13Hz. This sort of tweaking is easy to do with a signal generator and oscilloscope; ‘posh’ analysers such as the Audio Precision don’t go down low enough. Distressing distortion? Of course, the AP is very useful for checking the distortion, which is where some intractable problems were revealed. The Baxandall balanced input depends on a low-impedance virtual ground on the inverting input to work. Unfortunately, the input DC blocking capacitor (C1) and the LF step network raise the impedance at low frequencies, which causes the distortion to increase rapidly below 200Hz, as shown in Fig.1. It was still much better than connecting the transformer in the normal way, though. Subjectively, the effect was enhancing rather than destructive, especially on electro jazz pop, such as Level 42 (“you old codger”, I bet you’re thinking!). Switch off or carry on? The utility of a fully floating (Earthisolated) transformer input cannot be understated. I found the switch-mode power supply buzz and monitor noise from my desktop computer to be completely eliminated when being used as a sound source. With a stereo amp using a single power supply, the left and right amplifier signal Earths have to be joined together at the input jack. This causes an Earth loop, which invites hum. Putting ground lift resistors in can fix this, but distortion then increases. Using transformers breaks the loop. The amp also excelled when used as mid-range and tweeter amps in an active speaker system with a common power supply for all three amplifiers. Floating differential inputs greatly simplify Earth loop problems. The LF distortion was a long way from the required bandwidth, so it had no effect. Building bridges Until the boards arrive, I can’t test the transformer bridge-mode version of the amp, so it’s time for another diversion and back to Blumlein. Normally, when making a bridging driver, a circuit that gives two anti-phase outputs is used, such as two op amps, wired as an inverter and buffer. Very boring; I’ve already deployed 100 NE5532s in the last year. 0.5 Total Harmonic Distortion (%) I 'm waiting for a delivery from JLCPCB near Hong Kong. When the PCBs (designed by Mike Grindle) arrive, we’ll have an efficient way of building the cascode-input power amplifier and its unusual power supply, described in the July issue. Meanwhile, I can report results of some further investigations into the design. 0.2 0.1 0.05 0.02 0.01 .005 .002 .001 .0005 .0002 .0001 20 50 100 200 500 1k Frequency (Hz) 2k 5k 10k 20k Fig.1: the distortion curve with a balancing transformer added to the cascode power amp. Note the rise in LF distortion. The output is 15V peak-to-peak into 8Ω. Practical Electronics | September | 2025 R9 3.9kΩ +40V +10V +10V +30V +30V C2 47pF C3 4.7µF 35V Output 1 inverting Output 2 non-inverting C4 4.7µF 35V TR1/2 BC549C +18.5V +18.5V 0.7V + R7 120kΩ + C1 330nF Input R3 10kΩ +50V 2.5mA + R1 120kΩ R5 10kΩ 0.7V R2 100kΩ R4 1.5kΩ +1.5V +1.5V R6 1.5kΩ C6 100µF 50V 1mA +16.5V 1mA R8 100kΩ C5 100nF 2mA 2mA CRD 0V Fig.2: the LTP phase-splitter seems to have been forgotten. It can have single-ended or balanced inputs. It’s an ideal application for Blumlein’s major legacy, the long-tailed pair. A circuit that gives two anti-phase outputs from a single input used to be commonly known as a phase splitter. These were an important stage in valve amplifiers, where they drove the two output valves in push-pull. The classic circuit still used today is the Mullard 510, which used a long-tailed pair built around a double-triode ECC83 valve. I thought it would be useful to design a transistor version to add to my discrete circuit armoury. Older amplifiers used a single valve or transistor to accomplish phase splitting, called a concertina phase splitter. This had the disadvantage of the load on one output affecting the other output. The LTP phase splitter doesn’t suffer from this defect. LTP phase splitter The circuit is shown in Fig.2. The first version I built set the tail current Photo 1: the outputs of a phase splitter are the same amplitude. with an 8.2kΩ resistor, textbook style, which was a bad idea; not all the signal was transferred to the second transistor, some was lost in the resistor. This resulted in the second output being 20% lower than the first. The whole point of a phase-splitter is equal balance. The horrid sound and inefficiency of one amplifier in a bridge clipping, while the other amplifier is clean, defeats the whole point of the system. If the tail resistor is replaced by a 2mA current regulating diode (CRD), as shown in Photo 1, the imbalance is fixed. Alternatively, a constant current sink built around a JFET or bipolar transistor could be used. The op amp phase-splitter has the advantage of being able to drive low impedances, down to 1kΩ or so, with insignificant distortion. This phasesplitter has a high output impedance, defined by the collector resistors R3 and R5. Replacing these resistors with a current mirror would double the gain Total Harmonic Distortion (%) 0.5 0.2 0.1 0.05 0.02 0.01 .005 .002 .001 .0005 .0002 .0001 20 50 100 200 500 1k Frequency (Hz) 2k 5k 10k 20k Fig.3: the LTP phase splitter distortion with a 20V peak-to-peak output (unloaded). It’s quite high, but is essentially innocuous third harmonic at a consistent level. Practical Electronics | September | 2025 and output current capability for the same supply current. I measured the unloaded distortion with an 8V peak-to-peak signal at 0.03%, as shown in Fig.3. It was mainly third-harmonic, arising from the start of the symmetrical soft clip curvature. Such distortion is very innocuous and similar to that produced by speaker excursion limiting and tape saturation. Interestingly, the distortion did not increase with 600Ω loading, although the output voltage dropped. One thing I’ve noticed with differential operation is that the distortion of tantalum coupling capacitors cancels out because it is equivalent to operating two capacitors wired back-to-back. The distortion is further minimised in single-rail designs such as this, because the capacitors are polarised. So there is no characteristic bass tip-up in the distortion curve. The gain of the circuit for each output is 3.5 times, defined by the emitter resistors along with the collector loading. With differential outputs, there is a further gain doubling because one output goes up as the other goes down. The frequency response measures -1dB at 20Hz and 20kHz. The output rolls off around 300kHz, with output 2 dropping first because of the extra losses passing through an extra transistor. The capacitance of the constant current sink also has an effect. Maybe the long-forgotten Baxandall bootstrapped complementary constant current circuit would work well here, but that’s another investigation. Alternatively, I could design a cascode phase-splitter that needs no current sink at all. A practical, breadboarded 71 version of the phase-splitter from Fig.2 is shown in Photo 2. Bridging amplifier impedances Photo 2: a breadboard version of the LTP phase-splitter circuit shown in Fig.2. c be R3 C1 0V C2 Input R1 C7 R8 C6 Speaker + Speaker Gnd R12 R13 R11 C9 C3 R17 C11 + C10 TR4 R14 LED2 ZD1 + + TR5 TR7 b c e TR6 R19 C13 c be R15 R25 Dirty Gnd IC1 R10 LED1 + VR2 R21 R22 Clean Gnd R9 R6 C8 L1 C2 + C4 R23 R24 Speaker – C5 TR2 VR1 TR3 R20 R7 c be b c e R2 R5 TR1 R4 R16 R18 + TR9 TR8 Metal tab e c b e c b 54V Unreg 50V Reg From transformer 36V AC AC CLTP - POWER SUPPLY +50V Reg1 TR2 + C6 e c + R 6 Metal tab e b c TR1 C5 +54V Unreg1 AC b C4 +50V Reg Mains E D4 C7 R 2 + C2 F1 R1 D1 D2 R R 5 4 R 3 D5 +54V Unreg D3 D9 D8 + C3 72 Speaker Gnd1 Speaker Gnd2 Dirty Gnd1 Dirty Gnd2 Clean Gnd1 Clean Gnd2 Fig.5: the power supply board overlay. Note that sufficient pins are provided for stereo use. + C1 D7 D6 COMP-LT PAIR POWER AMP Fig.4: the cascode amplifier PCB overlay. The voltage loss on the discrete LTP phase splitter outputs could be reduced by making the power amplifiers non-inverting, giving a higher input impedance, thereby reducing the loading. This would be quite tricky with the PCB given here. One of the great advantages of bridging is that the output voltage is double that of a single amplifier. For a given speaker impedance, say 8Ω, the output power is theoretically quadrupled. This is because each amplifier has to deliver twice the current as well as there being twice as many outputs. The problem is that the speaker impedance ‘seen’ by the system is halved, ie, our 8Ω speaker looks like a 4Ω one (that’s why the current doubles). This doubling of current can stress the amplifiers, increasing the distortion and dissipation. One way around this is to reduce the transformer voltage a bit and double the VAS stage (TR3) current. In my case, this is great because I have loads of 15-0-15V 60VA toroidal transformers in stock. These should give around 50W RMS into 8Ω bridged for short periods with no stress. Another single amplifier could also be run for a tweeter amplifier, forming the basis of an excellent active speaker system. When using two amplifiers in bridge mode, it is important to use a single joint power supply; strange distortioninducing Earth loops can occur if separate power supplies are used with a common Earth. An interesting effect of using ClassAB amplifiers in bridge mode is that the current pulses drawn from the supply are full-wave rectified versions of the signal rather than half-wave. In theory, this means that the smoothing capacitors have an easier time because there is an effective doubling of the frequency of the signal-derived ripple on the supply rail. There is an advantage in dual-rail bridge systems in that no speaker current goes into the Earth; it’s entirely confined to the power rails. In this case, where the amplifiers are single-rail, the ‘negative’ rail and speaker Earth are one and the same. I suspect there is little advantage in practice, because supply wiring impedances are going to be more significant. Ding dong; the boards have arrived, so I can get the soldering iron out at last. Still, the LTP phase-splitter is a useful circuit block. Did you know it can form the basis of an all-pass filter or phase-shifter? Now there’s an interesting idea. Practical Electronics | September | 2025 Photo 4: the power supply board (C6 has been increased to 1000μF). The odd shape may provide space for a transformer. Building the cascode power amp The PCB overlays for the amplifier and power supply are shown in Figs.4 & 5. When initiating component insertion, remember that putting electrolytic capacitors in the wrong way is more catastrophic in single-rail circuitry because of the full polarisation voltage. Reversal of the output electrolytic, C10, will cause a big explosion and take your loudspeaker and output transistors with it! Positive is denoted by the square pad on the PCB. The same goes for the LEDs. The components are spaced out on the boards for educational use and experimentation, as shown in Photos 3 & 4. It is a good idea to space some resistors a few millimetres above the board Photo 3: the completed cascode circuit board with small heatsinks. Practical Electronics | September | 2025 Photo 5: it’s a good idea to space the resistors a few millimetres above the board, even if they normally run cool. (Photo 5) in case they burn up during fault conditions, causing damage such as holes in the PCB and possible fires. The Zobel resistor (R25), constantcurrent source resistor (R14), VAS emitter resistor (R24), driver transistor resistors (R16 & R17), output transistor resistors (R18 & R19) should be raised in all power amps for this reason. It’s quite reliable to do this on boards with plated-through holes (eg, doublesided PCBs). However on single-sided PCBs, it can result in broken pads and tracks if the components get pushed down, unless the leads are supported with special crimps or ceramic beads. The sealed Bourns-pattern TO5 trimmer resistors have very small holes; you can’t ram normal skeleton presets in. These tend to crack up and oxidise, anyway. A transistor ‘sandwich’ This is not lunch but a thermal construction where the driver transistors are bolted either side of the thermal sense (VBE) transistor, TR5, as shown in Photo 6. Instead of butter, thermal paste is used to ensure good coupling. They are held together using a 12mm-long M3 machine screw with a lock nut. Note that the metal tabs all face the same way; left, towards the nut/output capacitor. No insulating washers are needed because the inside of the device holes are insulated. I had changed the original Zetex transistors used for the more common BD139/40 devices for the drivers, TR6 & TR7. TR5 can be any NPN TO-126 device. I used a BD135 because it was at hand. I thought the original copper foil Photo 6: the ‘transistor sandwich’. I thought it was the best thing in thermal compensation, but there were side effects. 73 amp are made, it’s essential to check for oscillations with a scope on the output. Because there is less high-frequency open-loop gain due to the increased compensation, the HF distortion is worsened slightly from 0.02% to 0.03%, as shown in Fig.6. This could be fixed by using second-order compensation, but that’s for another investigation. I didn't notice any subjective differences. I’ve always agreed with Baxandall and Peter Walker (of Quad) that all competent amplifiers sound very similar up to overload. Simplifications Photo 7: soldering the original ZTX transistors to the PCB. This isn’t ideal, so I am designing a different board. mounting arrangement was too tricky. If you want to use the original transistors, they can be fitted as shown in Photo 7, but because they are centre-base rather than centre-collector, a bit of ‘leg crossing’ is called for. Component substitutions Transistor substitutions can be tricky with power amplifiers, and this proved to be no exception. Two component values had to be changed. Firstly, C8, the compensation capacitor, had to be increased to 56pF because the BD140 (TR7) is slower than the ZTX751, causing high-frequency oscillation. Also, the BD135 has a slightly lower (0.69V) switch-on voltage than the ZTX300 (0.78V) used for TR5. This meant that R12 had to be changed to 3.9kΩ. Interestingly, the ‘transistor sandwich’ actually over-compensates, the thermal coupling being too good. This could be beneficial if monolithic Darlington transistors were used for the output stage. Whenever substitutions on any power Output inductor L1 is only needed to prevent oscillation with capacitance from long twin-flex speaker leads. If the speaker leads are less than, say, 250mm (10 inches), such as in active speaker cabinets, it can be omitted and its position linked out on the PCB, along with resistor R21. Biasing If the board is to be used with a regulated driver supply, the bias chain can be replaced by resistors and regulator IC1 omitted. In this case, the zener diode supply still has a bit of hum. Remember that the bias input is a signal pin as well, so any noise on here is amplified. Heatsinking The output transistors are designed to be mounted onto a heatsink or a thick aluminium chassis. Insulating washers and bushes must be used since the metal tabs are at +23V. A thermal resistance no higher than about 3.4°C/W is needed. Power supply The main reservoir capacitor C1 should be 63V, not 50V as originally specified. With some mains transformers, the rail voltage can rise to 55V off load. The capacitor is unlikely to short out because Total Harmonic Distortion (%) 0.5 0.2 0.1 0.05 0.02 0.01 .005 .002 .001 .0005 .0002 .0001 20 50 100 200 500 1k Frequency (Hz) 2k 5k 10k 20k the electrode foil oxide is formed to the surge voltage rating (20% higher), but it’s best to design conservatively. Wiring The output transistors can be wired off-board with quite long leads, up to 200mm, before instability occurs. Note there are three Earth wires from the board. To keep impedances low, 16/0.2mm stranded wire is used for the high-current connections. The regulated supply and transistor base connections can be thinner, such as 7/0.1mm. The completed amplifier test jig is shown in Photo 8. Testing The first thing to do is put the trimmers in the correct position. The bias, VR1, should be set mid-way and the quiescent current, VR2, fully anticlockwise to maximum resistance. No speaker or load is ever connected with initial power amp testing. If you have a proper bench power supply that goes up to 50V (most only go up to 30V) with current limiting, one can take a cautious approach to testing with no blow-ups. If not, one can put 100Ω ¼W resistors in the power rails for protection. The smoke will soon tell you if there’s a fault! Remember that the output transistors can be un-powered by not connecting the 54V rail. This will enable low-level tests to be performed safely. DC tests Once the DC conditions are fine, with the output at somewhere around half-rail, the bias can be set to 20V with VR1. This can be checked across C5; the output emitter resistors (R18 and R19) are a good test point for this. If you want to check the VAS current source, there should be just over 1V across R14. The rail currents can next be checked with an ammeter connected in series with the power supply. The regulated 50V rail should draw around 7mA. The quiescent current can be set next, with an ammeter in series with the +54V rail. This should be zero until VR2 is advanced. There will be the usual point where it suddenly jumps up, so turn it slowly. Set it to around 100mA, and as the ‘transistor sandwich’ warms up, it will drop to the required value of 15-30mA. It will take around 10 minutes to settle down. This could be sped up by adding heatsinking, but that’s for the next board version. If you have a sensitive voltmeter, the voltage drop across R18 and R19 can be measured; it will only be a couple of millivolts. Fig.6: the distortion is slightly higher using BD139/40 driver transistors compared to the original ZTX751/ZTX651. The output is 12.7V peak-to-peak, 2.5W RMS into 8Ω. Signal testing 74 Practical Electronics | September | 2025 Now it’s time to get the signal generator out and give the amp a bit of a thrash. With no load, check it clips symmetrically and cleanly on a 1kHz sinewave. If it’s okay, try it at say 20Hz and 10kHz. Next, try square waves, looking for excessive ringing. Load testing The next test is to attach an 8Ω 30W load resistor. The output voltage will drop a couple of volts, but it should be possible to obtain over 40V peak-to-peak. VR1 can finally be trimmed to obtain symmetrical clipping. Only leave it powered for a minute or so at a time, or the resistor and heatsinks will get very hot. I got a slight increase in power to 28W/42.5V peak-to-peak compared to the Veroboard version because I used schottky diodes in the bridge rectifier rather than bog-standard ones. Bridge testing Now that we have two working boards, it will be interesting to explore bridgemode operation and the distortion effects of things like wiring layout. Initial tests were not promising, with distortion at HF more than doubling, confirming my experience that bridge-mode amplifiers generally sound inferior. It is possible to omit the output capacitors in a bridge amplifier, but do this at your peril, since both outputs are at +23V. A single short to ground and bang go the output transistors. I’m sure with further R&D, improvements will be made. Then we’ll build them into a nice case, but I’m out of time for now. PE Photo 8: the test jig at the prototype PCB stage – less precarious than the old Veroboard version. A pretty enclosure awaits. “You always say that”, says Kate, as she sweeps piles of mouldy circuits from the dining room table. 1591 ABS flame-retardant enclosures Learn more: hammondmfg.com/1591 uksales<at>hammfg.com • 01256 812812 Practical Electronics | September | 2025 75