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Teach-In 2026 by Mike Tooley World of Wireless – An Introduction to Radio and Wireless Technology Part 2 – Introducing RF components and circuits I n the first instalment of this series, we began by setting the historical stage, highlighting major milestones and scientific breakthroughs in radio before moving on to explain the propagation of radio signals. We provided an overview of the radio frequency (RF) spectrum, which spans from tens of kilohertz (kHz) up to hundreds of gigahertz (GHz), and examined how signals travel as ground waves at lower frequencies and sky waves at higher frequencies, with the aid of layers present within the ionosphere. Our Hands-On project involved the construction of a very simple medium-wave AM receiver for headphone reception. This month, we start by examining a simple radio frequency circuit before moving on to explore the principal components on which it is based. Our practical project features an improved portable medium-wave AM receiver. A VHF signal source To begin, let’s look at a simple RF circuit: a VHF signal source set to 70.4MHz, in the four-metre amateur band. The source delivers a stable, precise output a little below 10dBm (10mW) into a 50Ω load. It is ideal for antenna adjustment and receiver testing. Instead of generating the final 70.4MHz signal directly, we use an inexpensive quartz crystal oscillator at one-tenth that frequency and then multiply it by 10 in a frequency-multiplier stage. Fig.2.1 shows the circuit of the four-metre signal source circuit. It uses two 2N7000 N-channel Mosfet transistors (TR1 and TR2) as a fundamental-mode oscillator, and a third 2N7000 (TR3) as the ×10 frequency multiplier. Frequency accuracy comes from a 7.04MHz quartz crystal, easily sourced online. The series trimmer capacitor (TC1) allows fine adjustment to account for component and temperature variations. The tenth harmonic of the main oscillator is selected by means of a parallel-tuned circuit, L1 and C4, with the inductor made from five turns of tinned copper wire with the output coupling capacitor, C5, tapped at 1.5 turns from the ‘cold’ (i.e. supply) end. To permit tuning of the correct harmonic, L1 is fitted with an adjustable ferrite core. The three 2N7000 transistors are low-cost enhancement-mode Nchannel Mosfets (to be explained in more detail later). They are primarily designed for use in switching applications, but with appropriate bias arrangements, they can be used for linear as well as Class-C operation. They are suitable for output power levels of up to several hundred milliwatts. The whole circuit is powered by a 9V PP3 battery and enclosed in a screened die-cast metal enclosure. The output signal is shown in Fig.2.2 using a software-defined radio (SDR) receiver. Tuned circuits Tuned (or resonant) circuits are used to discriminate between signals at different frequencies. Fig.2.3 shows two basic configurations for a tuned circuit: series and parallel. The frequency at which both are resonant, f0, is given by the relationship f0 = 1 ÷ (2π√LC ). L L L L C C C C (a) Series (a) Series (b) Parallel (b) Parallel Fig.2.1: the circuit of our 70.4MHz signal source. Fig.2.3: ideal series and parallel resonant circuits (an ‘acceptor’ and ‘rejector’). 4 Practical Electronics | December | 2025 L L RLoss RLoss fo fo Frequency (Hz) Frequency (Hz) Frequency (Hz) Frequency (Hz) (a) Series (b) Parallel (a) Series Parallel circuits. Fig.2.5: the impedance-frequency characteristics of series- and(b) parallel-tuned circuitry to which it is connected) reduce its quality or Q-factor. Thus, as losses increase, the Q-factor is reduced, and vice versa. The Q-factor also determines the selectivity of a resonant circuit. A higher Q-factor is associated with narrower bandwidth and thus increased selectivity, whereas a lower Q-factor is associated with wider bandwidth and less selectivity, ie, Bandwidth = (f0 ÷ Q) Hz. For the series-tuned circuit shown in Fig.2.4, Q = (2πf0L) ÷ RLoss. Vout (mV) +V 33 L 23.3 C BF961 Vbias Vout Cout Vin Q = 23.6 L L RLoss RLoss fo fo Impedance Impedance (Ω)(Ω) All practical resonant circuits have losses, the most prevalent of which is that due to the series resistance associated with the winding of the inductor. Further losses occur at RF but, for simplicity, we will group these losses together and show them as a single series loss resistance, RLoss, as shown in Fig.2.4. The impedance-frequency characteristics of series and parallel resonant circuits are shown in Fig.2.5. It is important to note that the impedance of the series-tuned circuit falls to a very low value (ideally zero) at the resonant frequency; for a parallel-tuned circuit, it increases to a very high value (ideally infinite) at resonance. For this reason, series-tuned circuits are sometimes known as acceptor circuits, while paralleltuned circuits are sometimes called rejector circuits. The frequency response (voltage plotted against frequency) of a typical parallel-tuned circuit is shown in Fig.2.6. This characteristic shows how the signal developed across the circuit reaches a maximum at the resonant frequency, f0. The range of frequencies accepted by the circuit is normally defined in relation to the half-power (-3dB power) points. These points correspond to 70.7% of the maximum voltage, and the frequency range between these points is referred to as the -3dB bandwidth of the tuned circuit. Losses present in a tuned circuit (or imposed on it by the external Impedance Impedance (Ω)(Ω) Fig.2.2 a panoramic waterfall (spectrum, ie, power vs frequency) display of the 70.4MHz signal. C C C C 0.31 (a) Series (b) Parallel (a) Series (b) Parallel Fig.2.4: real tuned circuits have losses due to resistance and other factors. Practical Electronics | December | 2025 6.84 7.15 Frequency (MHz) Fig.2.6: the frequency response of a parallel-tuned circuit in a 7MHz RF amplifier. 5 C C C C L L L L L L L L C C C C 0 (b) Capacitively coupled (b) Capacitively coupled Fig.2.7: some coupled tuned circuits. (b) Capacitively coupled RF filters and selectivity A perennial(b)problem the Capacitivelywith coupled design of receivers and transmitters is the need to separate wanted signals from other signals on adjacent frequencies. We refer to this as selectivity. A single RF tuned circuit will normally exhibit a Q-factor between 50 and 110. Selectivity can be improved by using multiple tuned circuits, but this brings with it the problem of maintaining accurate tuning of each circuit. A band-pass filter can be constructed using two parallel-tuned circuits coupled inductively or capacitively, as shown in Fig.2.7. The frequency response of this type of filter depends on the degree of coupling between the two tuned circuits. Optimum results are obtained with a critical value of coupling (see Fig.2.8). Too great a degree of coupling results in a doublehumped response, while too little coupling results in a single peak in the response curve, accompanied Attenuation (dB) L L L L Un 40 de 60 fo r-c ou ple d Frequency (Hz) Fig.2.8: the frequency response of coupled tuned circuits. by a significant loss in signal. Critical coupling produces a relatively flat passband characteristic, with a reasonably steep fall-off on either side of the passband. Quartz crystals Quartz crystals provide us with an alternative to conventional L-C circuits and have several applications in RF circuits. Not only can they be used to accurately determine the frequency at which a circuit oscillates, but they can also be arranged to form highly selective filters. Quartz crystals usually consist of a thin slice of quartz (hexagonal crystalline SiO2) with film electrodes of gold or silver deposited onto opposite sides. A fine supporting and connecting wire is soldered at a nodal point on each electrode, and the complete assembly is enclosed in an evacuated glass or metal envelope. Lead-out pins or wires facilitate connection with external circuitry. The type of encapsulation, size, dimensions and pin spacing varies from one type of crystal to another. Fig.2.10 shows a selection of com- Fig.2.9: inductively-coupled tuned circuits in a VHF amplifier. The two trimmers are adjusted to get an optimal band-pass characteristic. 6 led C C C C up C C C C co erOv L L L L d Cc Cc 20 le coup cally c (a) Inductively Cc coupled Criti (a) Inductively coupled (a) Inductively coupled (a) Inductively C coupled monly available quartz crystals supplied in different encapsulations. The electrical equivalent circuit of a quartz crystal is shown in Fig.2.11. The device can be considered either series or parallel-tuned depending upon which of the two capacitors, Cs or Cp, is allowed to become dominant by external circuit conditions. The impedance/frequency characteristic of a typical quartz crystal is shown in Fig.2.12. Both series and parallel resonant peaks are evident; however, the frequency separation of the two is extremely small. At the series resonant point, the crystal behaves as a pure resistance, and any change in external capacitance will have little effect. At the parallel resonant point, the impedance of the crystal becomes inductive, and a change in the external circuit reactance will have the effect of ‘pulling’ the crystal frequency away from its natural parallel resonant frequency. Hence, if a crystal is to be used in a circuit that requires it to operate at parallel resonance, the load capacitance must be accurately specified. It is also worth remembering that the load capacitance is the dynamic capacitance of the total Fig.2.10: a selection of quartz crystals with resonant frequencies of 100kHz, 1MHz, 8MHz and 10MHz (left-to-right). Practical Electronics | December | 2025 Practical Electronics | December | 2025 Inductive Reactance CS L CP Parallel resonance 0 Frequency f R Capacitive Series resonance Fig.2.11: the electrical equivalent of a quartz crystal. There are two very close resonant frequencies: series and parallel. Attenuation AttenuationAttenuation Attenuation output, so it’s described as being more “active”. Consequently, quartz crystals with relatively low ESR values will require lower values of drive from the oscillator circuit to make the crystal resonate. Low-ESR crystals are therefore essential for lowvoltage/low-current oscillators. By introducing a combination of series and/or parallel reactance (both inductive and capacitive), the oscillation frequency of a crystal oscillator can be ‘pulled’ away from its nominal value by as much as ±0.5% of the nominal frequency (not necessarily symmetrically!). While this is normally an undesirable effect, it can occasionally be useful in providing a way to vary the operating frequency of a crystal oscillator over a narrow range. Some radio manufacturers have exploited this variable crystal oscillator (VXO) technique in incremental and finetuning controls. The frequency stability of a VXO arrangement is often significantly impaired at the edges of the ‘pulling’ range. Frequency stability can be preserved by keeping the excursion within reasonable limits. Attenuation Attenuation Attenuation Attenuation circuit measured across the crystal terminals. In parallel circuit design, the load capacitance should be selected to operate the crystal at a stable point on its frequency characteristic (usually as close to the series resonant frequency as possible). Quartz crystals exhibit extremely high Q-factors, which are many times larger than those that can be obtained with even the very best L-C tuned circuits. The reason for this is that the ratio of equivalent inductance, L, to series loss resistance, R, is exceptionally high. Crystals manufactured for fundamental operation are designed to oscillate at their basic resonant frequency, whereas those intended for overtone operation oscillate at, or very near, an integral multiple of their fundamental resonant frequency. Generally, the third overtone is preferred, although fifth, seventh, and even ninth overtone devices are available. At high frequencies, crystals become extremely thin, and are consequently more difficult and more expensive to manufacture. Thus, fundamental crystals are normally used at frequencies up to about 20MHz. Beyond this, overtone units are usually specified. The circuit load condition must be known to ensure the correct mode of operation (series or parallel). In series mode, the quartz resonator is operated in a low-impedance state. In the latter, it is operated in a highimpedance parallel-mode state. For parallel resonant crystals operating at high fundamental frequencies, it is critical that the load capacitance is accurately specified, usually in the range of 12pF to 30pF. In parallel mode operation, the crystal will operate with an impedance in which the inductive reactance is dominant. In this condition, the shunt capacitance is the principal external influence in determining the operating frequency of the oscillator. Part of the actual load capacitance consists of circuit and stray capacitances. These factors should be carefully evaluated to ensure that the actual load capacitance presented to the crystal is identical to the design value of load capacitance for which it is to operate. Like capacitors, quartz crystals exhibit an equivalent series resistance (ESR). This is the series resistance value that the crystal exhibits when resonant. A crystal with low ESR is easier to excite into oscillation and delivers a cleaner, stronger Crystal filters While individual quartz crystals are used as the frequency-­ determining element in Fig.2.12: the frequency characteristic of a quartz crystal. crystal-controlled oscillators, multiple crystals can be used to produce highly selective filters. By careful selection of resonant frequencies, it is possible to produce a network of crystals with near-perfect filter characteristics: a very low loss in the passband, coupled with a very high attenuation in the stopband. Fig.2.13 shows how the frequency response of a crystal filter compares with that of a conventional L-C circuit filter. Note that the response curve of the crystal filter shows a Bandwidth Bandwidth Bandwidth Bandwidth (a) L-C filter (a) L-C filter (a) L-C filter (a) L-C filter Frequency Frequency Frequency Frequency Bandwidth Bandwidth Bandwidth Bandwidth Frequency Frequency (b) Crystal filter (b) Crystal filter Fig.2.13: a comparison of L-C and crystal filters. Frequency (b) Crystal filter (b) Crystal filter Frequency 7 Fig.2.14: crystal filter frequency response. flatter top and steeper sides. This characteristic is essential for effective rejection of signals on adjacent frequencies. A typical crystal filter frequency response (showing the passband and stopband regions) is shown in Fig.2.14. The stopband attenuation for this filter is typically more than 60dB with a passband ripple under 3dB. The shape factor for the filter is usually specified in terms of the passband bandwidth (A) and the attenuation in the stopband, bandwidth (B). Thus, for a given value of attenuation, Shape factor = (Bandwidth B) ÷ (Bandwidth A). Typical values of shape factor range from about 1.5 to 3. The spurious responses (see Fig.2.14) associated with quartz Fig.2.19: these three ten-turn toroidal inductors have very different characteristics due to the different core materials. They exhibit inductances (left to right) of 265µH, 9µH, and 2.5µH. Only the right-most core is suitable for high-Q applications. crystal filters can be made less significant by using conventional tuned-circuit matching transformers. A typical 9MHz crystal filter for use in an HF communications receiver might have the following specifications: • Centre frequency: 9MHz • Passband bandwidth: 4kHz <at> -6dB • Stopband bandwidth: 12kHz <at> -60dB • Maximum ripple: 2dB • Maximum insertion loss: 3dB • Input/output impedance: 1.5kΩ • 60dB shape factor: 3 Unfortunately, good-quality crystal filters can be extremely expensive. For less critical applications, ceramic filters make a more cost-effective alternative. These are available in several standard frequencies, including 455kHz and 10.7MHz. We will be exploring their use later in this series. RF inductors and transformers RF inductors and transformers, like those shown in Fig.2.15, are frequently needed in RF circuits. For convenience, we’ve mainly used standard fixed-value inductors in our Hands-On projects, but you may occasionally need a specific value inductance that’s unavailable as a standard component or that needs to be made adjustable. In such cases, winding your own coil on a small former can be the best solution. The problem of determining the dimensions and current number of turns can be solved in various ways. However, coil winding is not a precise science, Fig.2.15: a selection of typical RF inductors and transformers. 8 Practical Electronics | December | 2025 Fig.2.17: the 9µH adjustable inductor. so experimentation will often be required. The coil winding chart shown in Fig.2.16 provides a quick rule-ofthumb method of determining the length of a coil winding required on a typical 7mm former. When supplied in insulated enamelled copper form, the three wire gauges quoted (26 SWG, 34 SWG and 40 SWG) will have typical diameters of 0.5mm, 0.3mm and 0.2mm, respectively. The example in Fig.2.16 shows how a 9µH inductor designed for operation at 9MHz and tuned by a 33pF capacitor will need to span Fig.2.16: a coil winding chart for 7mm formers. around 12mm of a 7mm former when wound with 34 SWG (0.3mm diameter) enamelled copper wire (ECW). The resulting inductor is shown in Fig.2.17. When using an adjustable ferrite core, the winding length (and number of turns) will typically need to be reduced by about 30%. The component shown in Fig.2.17 exhibits an inductance of between 7µH and 22µH over the adjustment range of Fig.2.18: using the Coil64 freeware application for inductor design. Practical Electronics | December | 2025 the ferrite core, corresponding to resonant frequencies of 6-10MHz when tuned with 33pF. Coil64 (see Fig.2.18) offers a very neat solution for designing many types of inductors. It is a 64-bit, cross-platform, open-source application licensed under GPLv3. It runs on desktop Linux, macOS (64-bit), and Windows (32/64bit) systems, and the application can be freely downloaded from https://coil32.net Toroidal ferrite cores are widely used in RF circuits for low-Q inductors, filters and impedance matching. Their core material determines critical characteristics such as inductance value, Q-factor and suitability for specific applications. One significant advantage of, and reason for using, toroidal cores is that their magnetic fields are mostly contained within and immediately around the core, so they interact less with other nearby coils compared to simple cylindrical forms. As illustrated in Fig.2.19, toroidal cores that have identical windings can yield vastly different inductance values depending on the material. One may produce high inductance (eg, 265µH), another moderate (eg, 9µH), and a third, low (eg, 2.5µH). 9 Fig.2.20: the RF Toroidal Core application, showing the frequency characteristic of a 45µH inductor from 100kHz to 20MHz (the orange trace; note the legend above the plot). There is a progressive reduction in inductance above 2MHz. Only certain materials are suitable for high-Q applications such as precise RF filtering and tuning. Therefore, selecting the appropriate core material is essential—some are optimised for low-Q filtering and matching, while others are designed for high-Q, high-frequency performance. Careful attention to core material properties is needed to ensure reliable and effective RF performance. The design of toroidal core inductors is greatly simplified with the use of software. Fig.2.20 shows Miguel Vaca’s excellent online toroidal calculator. You must specify the type and material of the core before experimenting with other parameters, such as wire gauge, turns and frequency. RF Toroid Calculator is available at: https://miguelvaca. github.io/vk3cpu/toroid.html With toroidal core inductors, it is important to be aware that inductance increases with frequency and DC bias. Power loss also increases with frequency and flux density, and this can be an important consideration when toroidal cores are to be used in power amplifiers. Finally, toroidal cores are often supplied with blue, grey, yellow, silver, red or green markings, but there appears to be no universally accepted standard for colour coding. One reliable supplier of cores, DX Engineering, uses the coding shown in Table 2.1. RF semiconductors The choice of semiconductor devices can be crucial in many RF applications. Devices designed primarily for low-frequency applications (below 3MHz) will usually not perform well at HF, VHF and UHF. Selecting the right device for the task can be crucial, so here’s a shortlist of selected RF semiconductors, including many of those that feature in our Hands-On projects. Fig.2.21: a selection of RF semiconductors used in this Teach-In series. Left to right: PN2222, BFR90, 2N7000, BF961, PGA-103, BAT42 and 1SV149. 10 Practical Electronics | December | 2025 Table 2.1 – Colour coding of toroidal cores supplied by DX Engineering Colour marking Core material Frequency range Application Blue 31 1-300MHz HF and VHF Silver 43 25-300MHz VHF and RF AGC (automatic gain control) circuits. The device is supplied in a four-lead TO-5 package, seen in the middle of Fig.2.21. A depletion-mode Mosfet is on until biased off, while the more common enhancementmode type is off until biased on. PGA-103 low-noise amplifier (LNA) Orange 52 200-1000MHz VHF and UHF Red 61 200-2000MHz VHF and UHF Green 75 0.15-30MHz MF and HF Pink 77 0.1-50MHz LF, MF and HF 2N2222 NPN silicon transistor The 2N2222 is a popular NPN bipolar junction transistor (BJT) used in RF and switching circuits. It offers moderate gain, fast switching speed and the ability to handle relatively high currents and voltages. Typically, the 2N2222 can handle collector currents up to 600mA and voltages up to 40V, making it ideal for driving inductive loads. The TO-18 metal case variant is still available, but the low-cost TO-92 packaged version is available as the PN2222. The device has a typical transition frequency (fT) of around 250MHz, so is suitable for operation up to and beyond 100MHz. This makes it ideal for non-critical HF and low-VHF applications. BFR90 NPN silicon transistor The BFR90 is a high-frequency NPN silicon BJT specifically designed for VHF RF applications. It has a low noise figure, together with excellent gain, making it ideal for use in amplifiers, oscillators and mixers operating up to 450MHz. The BFR90 comes in a TO-50 package, visible in Fig.2.21, with a maximum collector current of around 25mA and a maximum collector-emitter voltage of 15V. BF245C N-channel JFET The BF245 is an N-channel junction field-effect transistor (JFET) commonly used in a wide range of high-frequency, low-noise applications, including RF amplifiers, oscillators, and mixers operating up to Practical Electronics | December | 2025 150MHz. Variants like the BF245A, BF245B and BF245C have slightly different characteristics and may require different circuit parameters. The device is supplied in a TO-92 package. 2N7000 N-channel Mosfet The 2N7000 is a widely used Nchannel enhancement-mode Mosfet offering fast switching speed, relatively low on-resistance (typically less than 5Ω) and a very high input impedance (in common-source mode). The device is supplied in a TO92 plastic package and is capable of handling drain currents up to 200mA and drain-source voltages of up to 60V. RF applications include oscillators, buffer amplifiers, and drivers in low-power transmitters. The 2N7002 is a similar device in an SMD (SOT-23) package and comes in a few different variants (2N7002P, 2N7002K etc), although some of those are not as good for RF applications. BF961 dual gate Mosfet The BF961 is a dual-gate Nchannel depletion-mode Mosfet, well-regarded for its performance in high-frequency RF applications. The device is often found in the front-end stages of VHF and UHF receivers, where its excellent gain, low noise figure and high input impedance are particularly useful. The dual-gate configuration permits effective gain control and improved signal selectivity, making it ideal for use in mixers, amplifiers, Like the immensely popular SPF5189Z LNA, the PGA-103 is a high-gain, ultra-low-noise RF amplifier that offers outstanding performance across a range of frequencies from 560MHz to 3GHz. The device boasts a typical noise figure of around 0.5dB at 150MHz with a gain of 24-25dB. This makes it suitable for front-end RF amplifiers, satellite receivers and RF measuring instruments. The device has straightforward biasing requirements and exhibits excellent linearity, with a high third-order intercept point (IP3) of around 36dB, indicating excellent linearity. The PGA-103 operates from a 5V DC supply and comes in an SOT-89 surface-mount package. The PGA-103 distinguishes itself from standard general-purpose RF amplifiers like the BFR90 or BF961 previously described by offering a much lower noise figure and higher gain, particularly at VHF and UHF frequencies. While the BFR90 is suitable for small-signal amplification, and the BF961 excels as a dual-gate Mosfet mixer or AGC amplifier, devices like the PGA-103 or SPF5189 are better when low noise, high gain, and exceptional frequency response is required. BAT42 schottky diode The BAT42 is a small-signal schottky fast-switching diode that exhibits a low forward voltage drop of around 0.3V. The device is widely used in applications such as highfrequency rectification, clamping and protection circuits, especially where efficiency and fast response are critical. Its low capacitance and high reliability make it well suited to a wide variety of RF, digital and signal processing tasks. Schottky diodes conduct using only majority carriers, so they don’t suffer from the minority-carrier charge storage of standard PN diodes. This eliminates the reverserecovery delay when switching from forward to reverse bias. 1SV149 variable capacitance diode The 1SV149 is a variable capacitance diode, or varactor, designed 11 e L1 200μH (see text) R3 1kΩ R1 100kΩ P1-2 P1-1 R5 100kΩ IC1 TA7642 3 1 C2 100nF C1 10nF b R2 1kΩ C3 + VC1 270pF 2 C4 10μF + TR1 BC548 c +9V R4 1kΩ P2-1 + C5 10μF P2-3 To VR2 2.2μF D1 2.2V + P2-1 VR1 250kΩ 0V Fig.2.22: the RF module circuit for our new radio receiver. for electronic tuning applications. It offers a wide capacitance range that changes proportionally with applied reverse voltage, making it suitable for voltage-controlled oscillators, frequency modulators and tuneable filters in a wide variety of RF circuits. It offers low series resistance and stable performance across a wide range of frequencies, making it a preferred choice for both com- munications and general-purpose radio projects. Hands-On: A portable AM receiver Figs.2.24 & 2.25: the component layout (top) and track view (below) for the RF module. This month’s Hands-On project is a portable AM receiver, which builds on last month’s project by offering an RF gain control and much improved audio output from a small loudspeaker. The receiver uses a 9V power supply provided by a battery of six AA 1.5V cells connected in series (you could use a smaller 9V PP3 battery if you prefer). The 9V supply is necessary for the single-chip audio amplifier responsible for powering the loudspeaker. The receiver incorporates the same hand-wound coil and ferrite rod antenna that were used with last month’s practical project. The portable AM receiver uses two perforated copper stripboard modules, each measuring 25 × 64mm and arranged in nine strips each of 24 rows. One board is reserved for the RF circuitry (Fig.2.22), while the other is for the AF amplifier (Fig.2.23). As with last month’s project, tuning is managed by L1 and VC1, with L1 also serving as the antenna. L1 comprises 54 turns of 0.55mm ECW close-wound over a 70mm length of heatshrink sleeving, into which the ferrite rod is slid. The winding ends are secured with PVC tape. The resulting winding will have an inductance of around 200µH, and with the 270pF variable capacitor, will provide a tuning range extending from around 700kHz to 1.6MHz (see last month’s HandsOn feature for details). The precise value for VC1 is not critical, but a maximum value of 270-350pF will prove ideal. IC1 is a TA7642 radio receiver chip housed in a TO-92 package 12 Practical Electronics | December | 2025 S1 On/off +9V to RF board From RF board VR2 10kΩ P1-2 R1 47kΩ C2 10μF C6 + 10μF + 1 P1-1 + 6 8 C5 220μF 3 C3 10μF 7 4 + B1 9V 5 IC1 LM386N 2 C1 1nF P2-3 R2 10Ω C4 100nF P2-2 LS1 35Ω P2-1 0V to RF board Fig.2.23: the AF module circuit for our new radio receiver. + + + TA7642 BC548 GND collector 1 + + 1 2 3 2 IN base 3 OUT emitter + + D1 2.2V + 1 8 2 7 3 6 4 LM386N 5 Fig.2.28: the pin connections for the semiconductor devices used in our radio. Figs.2.26 & 2.27: the component layout (top) and track view (below) for the AF module. (see last month’s article for details). The supply to the TA7642 is regulated at around 1.5V by TR1 and D1. The AF module is based on a universally available LM386N 8-pin audio amplifier, IC2, in a DIL package. This chip produces reasonable audio quality at ample volume for domestic and portable listening. The component layouts for the two modules (viewed from the top) are shown in Figs.2.24 & 2.26, while the corresponding track layouts (viewed from below) are in Figs.2.25 & 2.27. The required track breaks can be made using a spot face cutter or small drill bit, and the links on the upper side of the boards are made using short lengths of tinned copper wire. The pin connections for the semiconductor devices are shown in Fig.2.28. When completed, the two stripboard modules, tuning components (L1 and VC1), volume control, RF gain control, loudspeaker, and battery holders can be mounted into an ABS enclosure of your choice, as shown in Fig.2.29. This can be the same enclosure used for last month’s Hands-On project. Connections to the off-board components (L1, VC1, VR1, VR2, etc) are made using the four 0.1-inch (2.54mm) pitch headers (shown as P1 and P2 on each board), together with short lengths of hook-up wire. Testing As always, it’s important to check the stripboard and internal wiring before applying power. When these checks are complete, insert six AA cells into the holders. Switch on and advance the volume Fig.2.29: the internal assembly and wiring detail of the radio. Practical Electronics | December | 2025 13 control. You should immediately hear some noise from the loudspeaker, and a quick sweep of VC1 should reveal several strong broadcast signals. If that’s not the case, switch off and recheck each board and wiring. For comparison, and to assist fault-finding, Table 2.2 provides the test voltages obtained from our prototype. Using it As with last month’s Hands-On project, our portable AM receiver is extremely sensitive and can be susceptible to local RF noise sources like computers, TVs and many other electronic devices. For best results, you will need to keep the ferrite antenna well away from other devices. The ferrite rod is directional; the strongest signal will be obtained when the antenna is side-on to the bearing of the station being received. These directional properties can also be advantageous, allowing you to null or minimise the effects of local noise and interference. Simply rotate the antenna until you find the position with the least interference and best signal! During the day, you will normally be able to receive several mediumwave broadcast signals at good strength. Noticeably more signals will be present during the hours of darkness. In the UK, BBC Radio 5 Live broadcasts on 693kHz and 909kHz nationally, and should be received at good strength, as will Lyca Radio (broadcasting on several MW frequencies from both London and Manchester). Live coverage of many sporting events as well as news and discussions is available from TalkSPORT on 1053kHz, 1089kHz, 1071kHz and 1107kHz. With the receiver placed in a favourable position, all these stations should be received easily and at good loudspeaker volume. Parts List – Improved MW Radio Receiver RF Module 1 25 × 64mm piece of stripboard (9 × 24 holes) 1 3-pin male 0.1in/2.54mm header (P1) 1 2-pin male 0.1in/2.54mm header (P2) Semiconductors 1 TA7642 single-chip radio receiver (IC1) 1 BC548 small-signal NPN transistor (TR1) 1 2.2V 300mW zener diode (D1) Capacitors 2 10µF 16V radial electrolytic (C4, C5) 1 2.2μF 25V radial electrolytic (C3) Resistors (all ¼W axial, 5% or better) 1 100kΩ (R1) 3 1kΩ (R2-R4) AF Module 1 25 × 64mm piece of stripboard (9 × 24 holes) 1 3-pin male 0.1in/2.54mm header (P1) 1 2-pin male 0.1in/2.54mm header (P2) Semiconductors 1 LM386N 8-pin DIL audio amplifier (IC1) Capacitors 1 220µF 16V radial electrolytic (C5) 3 10µF 16V radial electrolytic (C2, C3, C6) 14 1 100nF 50V ceramic (C4) 1 1nF 50V ceramic (C1) Resistors (all ¼W axial, 5% or better) 1 47kΩ (R1) 1 10Ω (R2) Off-board components 1 ABS enclosure (optional; see text) 1 270-350pF miniature solid-dielectric variable capacitor (VC1) 1 200mm-long, 10mm diameter ferrite rod (L1) 1 35Ω miniature loudspeaker (LS1) 1 250kΩ linear carbon track potentiometer (VR1) 1 10kΩ logarithmic carbon track potentiometer (VR2) 2 3×AA cell battery holders 2 battery clip connectors 8 brass or nylon M3 × 10mm hex stand-off spacers 8 short M3 panhead machine screws Table 2.2 – Expected voltages Device IC1 TR1 Coming up! Part 3 of our Teach-In series will introduce signals and modulation while delving into basic arrangements for CW, AM and FM transmitters. We will also explore RF semiconductor devices and show how they can be used to construct fixed and variable frequency oscillators in our two Hands-On projects. PE Join us then! 1 100nF 50V ceramic (C2) 1 10nF 50V ceramic (C1) IC2 Pin Voltage 1 0V 2 1.3V 3 1.1V C 9.4V B 2.2V E 1.6V 1 1.3V 2 0V 3 0V 4 0V 5 4.8V 6 9.4V 7 4.8V 8 1.3V NEW! 5-year collections See page 65 for further details and other great back-issue offers. Purchase and download at: www.electronpublishing.com Practical Electronics | December | 2025