Fullscreen HTML5Med Res (??MB)HTML4

AUDIO OUT AUDIO OUT L R By Jake Rothman A complementary cascode amplifier I n the analog audio world, amplifier circuits mostly all look the same. That is because it’s a mature technology and most engineers have settled on recycling circuits from Douglas Self’s books, simply because they’re proven to give excellent results at a reasonable cost. Unfortunately, this gives little scope for creativity. I enjoy designing slightly obscure discrete circuits, especially for musical instruments, where they still come into their own. For big Hi-Fi audio amplifiers, it’s difficult to beat Self’s circuits without a further increase in complexity, such as Bob Cordell’s DH-220C Mosfet power amp. However, there is still a creative +V TR2 Input 2 Non-inverting Re TR1 Input 1 Inverting Output RL 0V Fig.1: a complementary (NPN & PNP) cascode. It can be considered similar to a standard long-tailed pair. While low levels of even harmonic distortion are not a subjective problem in audio, it will likely make the total harmonic distortion (THD) higher on the specification sheet. window available between applicationnote design, such as the TDA2050 chip, and Self’s “blameless” architecture. This is where the complementary cascode circuit comes in (Fig.1), along with other early transistor circuits from the 1960s. It offers the AC performance of the long-tailed pair (LTP) with only two transistors rather than five, if a current mirror and sink is included. This complementary cascode configuration could be considered a type of complementary LTP. The catch is it exhibits a DC offset of 1.2V, which is why it is rarely used. This is because the two base-emitter (VBE) drops add up rather than cancel. Having said that, fixed offsets don’t matter if the amplifier is configured as a single-rail AC-coupled design with an adjustable half-rail bias. The complementary LTP needs no constant-current circuit since there is no tail current to consider, because one emitter directly feeds the other. Also, only one emitter resistor (Re) is needed. With the values shown in Fig.3, the gain is reduced to around 2.2× from 100×, with a commensurate reduction in distortion and an increase in signal handling capability. The even-order distortion cancellation is not as good as with the LTP, though, since the pair matching of complementary transistors is never going to be as good as same polarity devices. Complementary cascode buffer Before designing a power amplifier using a new circuit topology, it makes sense to start with a simpler practical circuit, like a buffer. As with the LTP, most cascodes have two inputs, with one input often connected to a fixed bias voltage. With the complementary cascode, one input is inverting and the other non-inverting. As in op amp circuits, the inverting input is often used for negative feedback. Note that in this circuit, the voltage amplifier stage (VAS), a common emitter stage, is inverting. So the inputs to the cascode have to be flipped. To reduce the distortion of tantalum and electrolytic output coupling capacitors, negative feedback (NFB) can be used, as shown in the op amp circuit in Fig.2. The 22nF capacitor has to be a low-distortion film type. The circuit also needs to be loaded to avoid a very low-frequency hump in the response. This technique can also be applied to the complementary cascode buffer shown in Fig.3. A similar technique is used for the power amplifier speaker coupling capacitor. ½ NE5532 + TR2 BC549 Output – 100kΩ R2 68kΩ 0V ½ NE5532 10µF 25V + Output + Input C1 100nF Input – 100kΩ 100kΩ RLoad ≈2.2kΩ 0V 0V 22nF Fig.2: one of my favourite negativefeedback tricks I developed to avoid distortion from output electrolytic caps. Practical Electronics | July | 2025 R3 47kΩ 0V R8 10kΩ TR1 R1 BC559 470Ω 9.8V C2 + 6.8µF 16V C6 100nF 11.6V R5 330Ω R4 100kΩ +24V C5 22nF C3 22pF 12.6V 12V R6 330Ω 0.7V R12 820Ω R10 220kΩ TR4 BC337 R7 100Ω TR3 BC549 C4 10µF 25V + 10µF 25V + Input R11 47Ω Output 12mA R9 1kΩ 0V Fig.3: a buffer amplifier using a complementary cascode as a complementary long-tailed pair. I breadboarded this to prove the technique works. 57 Here’s a funny aside. I knew one engineer who inadvertently relied on the parasitic inductance of the big output electrolytic (C10) to avoid using an inductor on the output. He got his comeuppance when the capacitors were “improved” with a lower equivalent series resistance (ESR) and the amps oscillated and blew up. An interesting characteristic of the complementary cascode that I noticed is that it has less switch-on/off thumps compared to the LTP, which tends to ‘snap on’. Power on/off transient generation is an area of analog circuitry that has not been fully researched yet (perhaps this would be a good topic for someone’s PhD). As a designer of active loudspeaker systems with up to eight power amplifiers, I will do anything to get rid of thumps, thereby avoiding the unreliability of speaker protection relays. blocking capacitor is used on the output so that the half-rail bias does not appear as DC across the speaker. I’ve found these capacitors can cause problems when used with amplifiers above 40W RMS because they can generate significant distortion from the high current passing through them. With single-rail designs, the capacitor is polarised at half rail, so the distortion from polarity reversal does not occur. The output capacitor also gives reliable DC fault protection, and the output transistors are also protected from output short circuits at low volumes. This is because the negative feedback loop at DC isn’t broken. Of course, if the amplifier is playing full pelt when a short occurs, the capacitor offers no protection. One major problem with single-rail power amplifiers is that the signal ground 0V can become contaminated with the half-wave rectified currents from the lower output transistor. This means that for the same circuit, the THD is always higher. Greater care has to be taken with the wiring layout. If gain is required, the NFB can be reduced using a potential divider. The amount of gain for low distortion is limited in this circuit, because the open-loop gain is low compared to an op amp. However, the gain can be increased by a bootstrapped or constant-current collector load for resistor R9. The power amplifier The first commercial power amplifier using the complementary LTP I had was a Bi-Pre-Pak module; readers over 60 will remember them being advertised in a two-page spread in PE. The Sterling Sound SS125, designed by ex-Sinclair engineer Richard Torrens in 1977, inspired this circuit. It intrigued me because it had a 0.02% THD rating rather than the typical 0.1% or more at the time. I used it as a midrange driver amplifier, along with Ben Duncan’s Hi-Fi News active crossover in 1980, to make a very successful sound system. I was only 18 at the time, and we had great parties until my Dad got fed up. Loads of ladies who wanted to be medics turned up, but they paired up with those men destined to become dentists. Sadly, they got paid much more for drilling holes in teeth than I did for drilling PCBs. Bias generator Let’s now look at the features of the complementary cascode power amplifier circuit I’ve designed, shown in Fig.4. For low-level circuits such as preamps, a resistor-based potential divider to generate the half-rail bias is fine. For a Hi-Fi class AB power amp running on a typical unregulated supply, that is not good enough. This is because the inevitable power supply voltage fluctuations can be fed back into the amplifier via the bias voltage, causing it to modulate itself. Output inductor When feeding long loudspeaker leads, as in typical domestic Hi-Fi setups, the capacitance can cause power amplifiers to oscillate. A damped inductor network (L1, R21) is often used to isolate this. If the leads are short, say in a guitar amp combo or active speaker, it can be omitted. The Zobel network alone, C12 and R25, will suffice to maintain stability. A single supply rail The traditional LTP is perfect for dual-rail DC coupled-amplifiers, where the speaker is connected directly to the output. With single-rail amplifiers, a DC- +50V regulated 8.5mA R8 12kΩ LED1 Red 3.2mA 2.5mA 4.5mA R1 2.2kΩ C1 2.2µF R3 47kΩ TR1 BC546B R12 6.8kΩ 19.5V C6 470nF Bias TR2 BC556B point C4 2.2nF R6 820Ω + R7 4.7kΩ R23 330Ω C5 10µF 25V IC1 k TL431 Adj R9 27kΩ TR6 ZTX651 mal Ther link T lin her k ma l 21V R5 160Ω Iq adjust 23V output bias C9 6.8µF 6.3V Tant CW C8 10pF 15-37mA + VR2 5kΩ R10 2.2kΩ TR3 ZTX694B a TR7 ZTX751 R24 13Ω R17 100Ω ZD1 43V R2 10kΩ 58 C7 33pF R19 0.1Ω TR9 BD911* 4.5mA 0V R20 470kΩ C10 2200µF 50V LED2 Orange CW DC bias adjust R18 0.1Ω C11 220nF Clip VR1 5kΩ +54V unregulated (drops to +46.5V on full power) TR5 ZTX300 R11 5.6kΩ C2 470nF *On 6.8°C/W heatsink TR8 BD912* R16 100Ω R15 33Ω 0V Input 4.5mA R13 15kΩ 0.75mA C3 100nF R14 220Ω TR4 BC556B + R4 470Ω Clean ground DC negative feedback AC negative feedback Practical Electronics | July | 2025 This raises distortion and causes the speaker cone to flap about at low frequencies, an effect I call ‘bias pumping’. If you look at the data sheets for some power amp chips, it will give a higher distortion figure for single-rail use compared to dual-rail (unless it’s been measured with a stabilised bench power supply). The solution is to regulate the bias voltage with a zener diode. In this case, a TL431 ‘programmable zener’ IC is used because it is adjustable (with VR1). If a regulated supply is used for the input stage, a decoupled resistive divider will suffice. It’s always a good idea to design an amplifier, its power supply and the speaker to be used as one system. I trim the bias for symmetrical clip at the lowest dip in the impedance curve of the speaker. Do it quickly to avoid burning out the driver. Input and VAS power supply This might be considered an overthe-top upgrade, but it gives an audible difference to me. As I’m now 62, I can no longer hear the extreme highfrequency distortion harmonics that most complex power amplifier circuits aim to minimise. Despite that, I still measure high-frequency distortion. I’m also very aware of hum and its modulation effects. It’s very difficult to regulate the power supply to the whole amplifier; it’s often more complex than the amplifier itself, and it reduces the maximum output on peaks by 30%. Luckily, regulating the low-level input and voltage amplifier stage parts of the circuit is quite easy. In this design, the output noise and ripple was reduced from 10mV to 2mV by adding regulation for the early, low-power stages. Use of a more complex LM317 high-voltage regulator circuit rather than the simple zener scheme (ZD4, ZD5) used here will reduce it still further. If you want to add rail regulation to a dual-rail amplifier, two regulators will be L1 10µH C12 100nF 100V R25 4.7Ω 1W Fig.5: a suitable power supply for the amplifier, with a voltage doubler built around diodes D1 and D2 acting as a charge pump. TR2 pulls the regulated rail down quickly for fast muting on power-off. Output Mains input R22 2.2kΩ 0.5W LS1 8Ω F1 2A PTC fuse T1 60VA 18V 36V 230V R3 22kΩ D3 1N4148 C4 + 2.2µF 63V C2 22µF 100V 6A 200V Bridge L – Faster switch-off One of the problems with the traditional linear power supply setup is that it does not switch off quickly; it carries on working until the capacitors discharge. This can let through thumps from previous stages. By turning off the low-power regulated supply quickly, the power amp can be muted. This is accomplished by a discharge circuit (TR1 & TR2) that pulls down the rail when the AC from the mains transformer goes away. This is driven by its own rectifier diode, D3, with minimal smoothing provided by capacitor C4, so it powers down before the rest of the circuit. The power supply circuit is shown in Fig.5. An 18-0-18V 50-60VA transformer is more than sufficient for one channel. A top-notch approach to prevent Earth loops in a stereo amplifier, would be to use a 35V + 35V 100VA dual secondary R4 22kΩ 13V C5 470nF D2 1N4002 D1 1N4002 Dirty ground Fig.4: the complete power amplifier circuit. The currents through the driver transistors are about 4.5mA because the output transistors are not fully switched on under quiescent conditions. R3 and C2 form an LF step network to prevent the LF hump. Practical Electronics | July | 2025 78V R2 1kΩ 2.5W WW R6 220Ω 0.5W +50V regulated +54V unregulated N + C1 10,000µF 50V R1 1kΩ 0.5W TR1 BC337 + 18V Speaker ground TR2 BC639 or BD139 R5 10kΩ + R21 3.3Ω 1W series to indicate clipping. The power is still reduced from 30W RMS into 8Ω to 26W, though. Since low-power amplifiers, like this, will inevitably clip occasionally in real use, this modification is clearly audible and sounds excellent as a guitar amplifier. It also allows us to use most of the available voltage from a given power supply transformer. With the transformer being the most expensive component in any power amplifier, it more than pays for itself. The Hi-Fi enthusiast’s technique of using a 100W power amplifier to avoid dirty clips when driving small speakers seems a very expensive solution. needed, but with a single-rail amplifier, you need only one. The famous Quad 303 single-rail amplifier had full regulation, although it was mainly used to limit the maximum voltage on the output transistors. The complexity of regulation means it’s rarely done. (Dual regulated audio power amp supplies are available, such as those by Hypex, but being SMT switchmode designs with over 100 parts, I’ve found them to last only last five years and be uneconomic to repair. That is unacceptable in the recording industry.) Also, with current mirrors and other improvements to the input and driver circuitry, the common-mode rejection of the amplifier can be made very high. However, this only works until the output transistors saturate at clipping; then all the power supply noise is let through, giving very dirty-sounding clipping. This horrid effect can be avoided by clean-clipping the driver stage first, but this will again reduce the maximum output power on peaks by around 30%. This is because the clipping level must be set below the lowest voltage that the unregulated power supply sags to on kick drums and other peaks. As usual with electronics, there’s a nice solution: since the current requirements are only around 10mA, a simple voltage doubler can be used with a zener regulator, and we can then set the rail voltage a few volts higher than the output stage supply rail. A defined ‘clean soft clip’ level before the output stage can then be set by a zener diode (ZD1) with one LED in C3 + 220µF 100V D4 27V 1.3W + D5 24V 1.3W C6 100µF 63V Clean ground Loudspeaker ground Dirty ground E C7 10nF 59 toroidal transformer driving two separate PSU boards. Voltage amplification stage (VAS) This is just the standard configuration, a common-emitter stage with a dynamic load. The load here is a constant-current source, rather than a bootstrapped collector resistor, because it gives lower crossover distortion. Its higher voltage loss is obviated by use of the higher voltage rail from the voltage-doubled supply. One problem with the standard common-emitter VAS is that it is subjected to the full rail voltage swing between emitter and collector (VCE). With bad examples of some older types of transistors, such as the BFY51 and BC107, this could result in distortion from the Early effect. This is where the gain and capacitance of the transistor is modulated by its VCE voltage. I remember having to buy special Motorola transistors that avoided this problem. These were used in Bailey and Quad amplifiers. Another source of good audio transistors was Ferranti in Oldham, near Manchester. Their ZTX E-line series ‘Zetex’ transistors were used in the groundbreaking PE Gemini and Orion amplifiers. They are still in production by Diodes Incorporated 50 years later! 2A 60V ZTX651 and ZTX751 transistors are used in the driver stage here. The Orion VAS transistor was a Ferranti BFS61 or ZTX451. I have found the ultra-cheap BC546B only works okay for swings up to 30V. For this amplifier, I used a ZTX694B with a 13Ω emitter resistor, which exhibits little Early effect. It has a VCE rating of 120V. High-voltage transistors always have less Early effect, but often the Hfe is low; the ZTX694B has an Hfe of 400. Adding a small amount of emitter resistance (R24) also helps reduce distortion at high levels. Conversely, adding resistance increases distortion at low levels due to open-loop gain reduction. I did try a cascoded VAS stage based on the Gemini circuit, but I found it offered no improvement on the ZTX694B. D. S. Gibbs and I. M. Shaw, who designed both PE amplifiers, were Ferranti application engineers at the time. Shaw was also the inventor of a diode dodge to improve the symmetry of quasi-complementary output stages. Later, this was improved by Baxandall, then adopted commercially by Rogers and Naim. Low gain for low noise Most power amplifiers have too much voltage gain for use in active speakers; typically 20× to 40×. This boosts the noise from the active filters, creating a 60 Photo 1: a power amplifier and its supply on a bit of wood! A technique I often use for prototyping. Photo 2: a close-up of the driver transistor heatsinking and thermal coupling arrangement. A bit of thermal paste in the ‘sandwich’ helps. noticeable hiss. Peter Baxandall reduced the gain of the Quad 405 amplifiers while increasing the crossover level in the pioneering KEF KM1 active speaker, resulting in a system as quiet as a passive speaker. So with this amplifier, I also decided to go for a much lower gain. I’ve even made one unity-gain stable so you can use it as an active filter itself, or put an active gain control around it. I also made it inverting, which allows a fully filtered half-rail bias generator to feed the non-inverting input (base of TR2). Inverting amplifiers generally have more noise than their non-inverting counterparts, but this is only if you need a high input impedance. In this case, the input impedance is kept low, at 2.2kΩ; this is high enough for most op amps and reasonable discrete circuits to drive. The inverting configuration is also useful for bridging with another power amplifier and for adapting to a balanced input. Now we have something sufficiently different from other amplifier circuits to make the design work worthwhile. Output stage I’ve opted to use the complementary emitter follower pair (CFP) output configuration here since it works well for powers below 50W, where the output transistors can be fairly small, with low capacitance. For powers above this, and where devices need to be paralleled, I use the Darlington emitter follower (EF) topology, which gives lower distortion and has better HF stability. By Darlington, I don’t mean monolithic Darlington transistors, where there is a driver and output transistor on the same chip, since this exacerbates thermal instability. The driver transistors have to be separate from the output transistors to avoid being heated up by them. Philips used monolithic Darlingtons successfully in their active speakers by putting a thermistor in the V BE multiplier. For small amplifiers, the CFP gives Practical Electronics | July | 2025 Photo 3: the prototype stripboard is slightly messy because the board is too short. Photo 4: the star wiring to capacitor pins is visible on the power supply underside. lower quiescent dissipation. Also, the VBE multiplier transistor only needs to be in thermal contact with the drivers for thermal stability, avoiding the hassle of mounting it on the main heatsink. The crossover distortion of the CFP is more ‘spiky’ than the Darlington EF and occupies a smaller voltage swing. It also has poorer stored charge removal, hence the HF distortion is a bit higher at 10kHz. The collector load resistors on the drivers are quite important for low distortion at high levels. Generally, for amps of this size, 100Ω is optimum, setting the driver current to 4.5mA. The output transistors TR8 and TR9 are not critical. I’ve used old TIP41/2C, BD911/912 and even the venerable TIP3055/2955 complementary pairs. Of course, if you want lower HF distortion, you could use modern, faster types. Building and testing The complete prototype assembly is shown in Photo 1. Yes, it is a real “breadboard”; a nice metal enclosure is Practical Electronics | July | 2025 not warranted for an initial prototype (the mains wiring has to be well-insulated!). I always build prototype power amplifiers on stripboard; breadboards are just too prone to bad connections and short circuits, which could cause it to blow up. Normally, stripboard is only suitable for simple circuits, which is true if one is going to build it in one go. The key to building a complex, fragile circuit on stripboard is to build it in testable sections. As always, electronics can be easily partitioned into its constituent stages, and this is the key to its success as an engineering discipline. Try doing this when building a railway bridge, though, and life becomes much more difficult! Always use a bigger board than you need. If you run out of holes, you have to start soldering onto component leads, which quickly becomes messy. The board can always be snipped down to size with side-cutters (or cut with a hacksaw) later. Although a good PCB layout designer does not usually place components in the same physical position as the schematic, with strip board prototypes, it makes it easier to test if it’s wired like a circuit diagram. That is with the positive rail is at the top, the negative rail on the bottom and the signal flow from left to right. The stripboard is shown in Photo 3. If you have to move something to optimise the layout, always test it before adding something else. Once you have more than two errors, the time taken to get things working rises disproportionately. Add loops of tinned copper wire to attach test probes and crocodile clips. Veropins don’t work; the spring-loaded test leads can ping off. An extra-big loop should be used for the Earth, since there may be up to five leads attached. For this amp and many others, I’ve built and tested the circuit blocks in the following order. Always ramp up the voltage slowly using a bench power supply and set it to a low current limit, around 250mA: 1. Create the current source bias voltage using LED1. This tells you circuit is powered. 2. Add the current source, TR4. Make sure the current is correct; in this case, 4.5mA. 3. Add the input transistors, TR1, TR2 and VAS TR3. 4. Test the gain with feedback resistors R1 & R2 wired into the circuit. 5. Add VBE multiplier transistor TR5 and test that the VR2 preset works. Set the voltage across its collector and emitter to a minimum. 6. Add driver transistors TR6 and TR7. Since this is basically a push-pull emitter-follower, the feedback can be connected and a basic test performed. 7. Finally, add output transistors TR8 and TR9. Gradually increase the current limit on the supply. To thermally couple TR5, TR6 and TR7, sandwich VBE multiplier TR5 between the two drivers (TR6 and TR7), wrapping them all together with a copper strip (see Photo 2). This also acts as a low-power heatsink. I used another Zetex transistor (a cheap low-spec one) for TR5 because the flat E-line shape allows them to fit together with maximum surface coupling. Adjust VR1 for symmetrical clipping with a 50W 8Ω load resistor and VR2 for the lowest distortion at low powers (3V peak-to-peak). This will typically give a quiescent current of 15-37mA. The great thing about getting a stripboard circuit working is that you can guarantee the next developmental stage, a proper PCB, will work better. It will have lower parasitic capacitance and resistance. Power supply construction Stripboard is less than ideal for building power supplies since it is very 61 0.5 Total Harmonic Distortion (%) difficult to make star connections on capacitor pins as shown for C1 in Fig.5 This technique is necessary to reduce the hum induced by charging pulses and prevent noisy Earth currents from impinging on the audio. Power supplies also need low resistance tracks. It is possible to buy 0.2-inch (5.08mm) Veroboard, which is usable, but it’s rare. It’s best to use perfboard (which has no copper) with tinned copper wire. This technique can give better results than a PCB, probably because the tinned copper wire has a substantially larger cross-section than a PCB track with standard thickness. I occasionally resort to soldering tinned copper wire onto PCB tracks to lower their resistance. The underside is shown in Photo 4. 0.2 0.1 0.05 0.02 0.01 .005 .002 .001 .0005 .0002 .0001 20 50 100 200 500 1k Frequency (Hz) 2k 5k 10k 20k Fig.6: the distortion vs frequency curve for the power amplifier. This was done at low power (2W into 8Ω). Performance The distortion vs frequency curve of an amplifier is always revealing; this one is shown in Fig.6. The THD+N is around 0.003% midband at low powers, so the complementary LTP works almost as well as a traditional LTP. If a single input transistor stage had been used, the distortion would have been about 0.01%. There is little increase in distortion at 20Hz, which illustrates the effectiveness of NFB around the output capacitor. There is a rise to 0.02% at 10kHz at all levels, which is not so good. This suggests the amplifier is running out of open-loop gain at high frequencies. This is because the compensation, layout and output stage need further optimisation. The complementary LTP’s subtraction for the negative feedback also possibly doesn’t work so well at higher levels compared to a properly matched LTP. The LF/mid-band distortion of the amplifier steadily increased to 0.005% at 12.5W and to 0.018% at 20W, so we were back to the stated specification of the Stirling Audio module. However, there could be some other problems, such as in the component layout. I’ve effectively reinvented the wheel here, and it does not give a Douglas Self Photo 5: the power supply. Note the connections going directly to the pins of the big reservoir capacitor, C1. level of performance. However, it’s still worth adding the complementary LTP to the armoury of discrete building blocks, along with this small power amplifier system suitable for active speakers. These days, linear power amp chips are often single-sourced and expensive compared to a handful of discrete transistors, so if you have the time, roll PE your own amp! 1591 ABS flame-retardant enclosures Learn more: hammondmfg.com/1591 uksales<at>hammfg.com • 01256 812812 62 Practical Electronics | July | 2025 Parts List – Complementary Cascode Amp 1 6 × 5in (150 × 125mm) piece of stripboard 2 5kΩ trimpots (VR1, VR2) 1 10µH 5A suppression inductor (L1) 2 6.8°C/W flag heatsinks 1 10mm length of 0.5mm-thick copper strip (from builder’s merchants) various screws, nuts etc Semiconductors 1 TL431 programmable voltage reference IC, TO-92 (IC1) 1 BC546B 65V 100mA NPN transistor, TO-92 (TR1) 2 BC556B 65V 100mA PNP transistors, TO-92 (TR2, TR4) 1 ZTX694B 120V 1A NPN transistor, E-Line (TR3) 1 ZTX300/ZTX108 45V 100mA NPN transistor, E-Line (TR5) 1 ZTX651 60V 2A NPN transistor, E-Line (TR6) 1 ZTX751 60V 2A PNP transistor, E-Line (TR7) 1 BD912 100V 15V PNP transistor, TO-220 (TR8) 1 BD911 100V 15V NPN transistor, TO-220 (TR9) 1 3/5mm red LED (LED1) 1 3/5mm orange/amber LED (LED2) 1 43V 400mW zener diode (ZD1) [eg, BZY88C43V43V] 2 100Ω 1 33Ω 1 13Ω 1 4.7Ω 1W 1 3.3Ω 1W 2 0.1Ω 1-2.5W (wirewound) Power supply parts 1 4in × 2in (100 × 50mm) piece of perfboard 1 18-0-18V 50VA toroidal or 37V 1.6A EI-core transformer 1 2A 60V PTC thermistor (‘solid state fuse’) All 60 issues from Jan 2019 to Dec 2023 for just £49.95 PDF files ready for immediate download Q ty 1 1 1 2 2 1 1 1 1 1 2 2 1 1 1 2 1 2 1 1 1 1 Value 4-band code 5-band code 470kW 47kW 27kW 22kW 10kW 12kW 15kW 6.8kW 5.6kW 4.7kW 2.2kW 1.0kW 820W 470W 330W 220W 160W 100W 3 3W 1 3W 4.7W 3.3W Semiconductors 1 BC337 45V 800mA NPN transistor, TO-92 (TR1) 1 BD139 80V 1.5A NPN transistor, TO-225AA (TR2) 1 200V 6A bridge rectifier (BR1) 2 1N4004 400V 1A power diodes (D1, D2) 1 1N4148 75V 200mA signal diode (D3) 1 27V* 1.3W zener diode (ZD4) 1 24V* 1.3W zener diode (ZD5) * a single 51V zener could be used if available Capacitors 1 10,000µF 50V solder-tag electrolytic (C1) 1 220µF 100V electrolytic (C3) 1 100µF 63V electrolytic (C6) 1 22µF 100V electrolytic (C2) 1 2.2µF 63V electrolytic (C4) 1 470nF 63V polyester/MKT (C5) 1 10nF 50V X7R ceramic (C7) Resistors 2 22kΩ ±5% ¼W 1 10kΩ ±5% ¼W 1 1kΩ ±5% 2.5W wirewound 1 1kΩ ±5% ½W 1 220Ω ±5% ½W Practical Electronics | July | 2025 5-year collections 2019-2023 Capacitors (all ±20% tolerance unless noted) 1 2200µF 50V low-ESR electrolytic (C10) 1 220nF 50V X7R ceramic (C11) 1 10µF 25V electrolytic (Al or Ta) (C5) 1 100nF 100V polyester/MKT (C12) 1 6.8µF 6.3V tantalum (C9) 1 100nF 50V X7R ceramic (C3) 1 2.2µF 63V polyester/MKT (C1) 1 2.2nF 50V X7R ceramic (C4) 1 470nF 63V polyester/MKT (C2) 1 33pF ±5% NP0/C0G ceramic (C7) 1 470nF 50V X7R ceramic (C6) 1 10pF ±5% NP0/C0G ceramic (C8) Resistors (all ¼W ±5% or better) 1 470kΩ ±1% metal film 1 4.7kΩ 1 47kΩ 1 2.2kΩ ½W 1 27kΩ 2 2.2kΩ 1 15kΩ 1 820Ω 1 12kΩ 1 470Ω 1 10kΩ 1 330Ω 1 6.8kΩ 1 220Ω 1 5.6kΩ 1 160Ω NEW! 2018-2022 All 60 issues from Jan 2018 to Dec 2022 for just £49.95 PDF files ready for immediate download 2017-2021 All 60 issues from Jan 2017 to Dec 2021 for just £49.95 PDF files ready for immediate download 2016-2020 All 60 issues from Jan 2016 to Dec 2020 for just £44.95 PDF files ready for immediate download See page 33 for further details and other great back-issue offers. Purchase and download at: www.electronpublishing.com 63