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The completed prototype, highlighting
the construction of the output filter (L4
& C9). The positive lead is threaded
through a small toroid 5-6 times before
being soldered to the rear of the output
terminal. The capacitor is soldered
directly across the positive and negative
terminals as shown.
By LEONID LERNER
A regulated 125W HV
supply for valve amplifiers
Looking for a low-cost high-voltage (HV)
supply to run valve circuitry? Here’s how
to modify a PC power supply to produce a
700V or 400V HV rail.
V
ALVE CIRCUITS are not yet
dead. While transistors are
undoubtedly superior in most
applications, the valve still offers several unique advantages. This applies
first and foremost to its use in power
circuits.
There exists a substantial body of
opinion that valves outperform transistors in high-quality audio amplifiers,
especially in the power output stages.
The seriousness of these claims is
74 Silicon Chip
reflected in the fact that some very
reputable manufacturers offer valve
amplifiers at the top end of their
audio range. For the home constructor, reasonable-quality valve audio
amplifiers can be made for a modest
outlay using designs available freely
on the Internet. These amplifiers are
generally based on an EL34 or KT88
valve pair in the output stage, with
both valves being readily available in
Australia.
Another common application for
valves is in the output stages of RF
power amplifiers. They will operate
satisfactorily at frequencies of up to
about 30MHz, delivering up to 50W
per valve. Their main advantage over
RF power transistors, apart from being
somewhat cheaper, is that they are
much more tolerant of fault conditions.
When tuning a new power amplifier
design, parasitic oscillations are often
encountered which can easily destroy
expensive RF power transistors. The
valve, however, will live to see another
day. Valves are therefore much more
suitable for experimentation in new
designs.
Although valves are readily obtainable, one of the main problems in their
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exploitation is the lack of suitable
power supply transformers. Both the
EL34 and KT88 are rated at a maximum
plate voltage of 800V, with supply voltages in the order of 500-600V needed to
extract maximum power and linearity.
However, the only readily available
high-voltage power transformers are
isolating transformers, which deliver
240V, and magnetron transformers
from microwave ovens, which deliver
1500V or more.
Clearly, both of these are unsuitable
for our application.
The easiest way around this is to
modify the switchmode power supply of a personal computer (PC), as
explored in a previous issue of SILICON CHIP (October 2003). The older
AT power supplies are readily available and have now become a surplus
item. They are designed to produce
about 200-300W, which is in the right
ballpark for our application. For little
cost, they include a ready-made PC
board and almost all of the components we need for a HV switching
power supply.
Moreover, due to its high operating
frequency, the switchmode power supply offers much better regulation and
far less ripple than can be obtained
from a traditional valve power supply
based on 50Hz AC rectification and
smoothing.
Basic considerations
At first, it would appear that getting
a PC power supply circuit to operate
at high voltages involves just a few
changes to the procedure outlined in
the previous SILICON CHIP article. In
particular, the number of power transformer secondary turns would have to
be increased and all diodes, capacitors, and inductors would have to be
replaced with high-voltage types.
The resistive ladder used to sense
output voltage would also have to be
changed. However, after a few trials, I
found that the switching power transistors did not last long and it soon
became clear that getting the circuit to
Fig.1: the power section of the modified high-voltage supply. Using the
values shown, the output is a wellregulated 700V, suitable for driving
two power valves. You can also build
a 400V version by winding T1 & L2
accordingly and selecting alternate
values for capacitors C1 & C2 and the
R3-R5 divider string (see text).
siliconchip.com.au
July 2004 75
Fig.2: the schematic of a typical
control section based on the TL494
PWM controller. The only changes
needed here are the removal of the
over-voltage detection circuitry
and the addition of an over-current
indicator, based on Q7 and an LED.
operate at 700V would entail a more
substantial redesign.
The main problem is that the voltsper-turn ratio used in the secondary
winding of a standard PC ferrite-cored
transformer (operating in step-down
mode) is about one turn per volt output. This means that 700 secondary
turns would be required for an output
of 700V.
And that’s where we quickly run
into problems. The power handling
capacity of a coil, without considering
insulation, is almost directly proportional to its volume. For example, if
we wish to double the output voltage
produced by a transformer, we have
to double the number of secondary
turns, and thus the coil length. The
resistance of the coil will also approximately double.
However, if the coil is to deliver
the same power, the output current is
halved so that the coil’s “ohmic” (I2R)
losses are halved. To compensate for
this, we can halve the wire’s cross-sectional area so that the overall volume
occupied by the coil is unchanged.
Unfortunately, a multi-layered coil
operating at high voltages and frequencies requires insulation whose thickness increases roughly proportionally
to the voltage. As a result, our coil does
not follow the volume law.
In fact, it is almost impossible to fit a
700-turn winding with adequate insulation into the space available around
the core of a standard transformer.
The reason for the large number of
secondary turns is that the original
PC supply uses a full-wave centretapped rectifier configuration, which
requires twice the number of turns
of a full-wave non-centre-tapped
configuration. However, even a noncentre-tapped configuration causes
problems.
For a start, it is difficult to fit even
350 turns in the space available around
the core. Also, the bridge configuration
has no “cool” end of the secondary
winding, with both ends alternatively switched between ground and
maximum voltage. This means that
heavy-duty insulation needs to be used
76 Silicon Chip
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between the primary and secondary
windings.
Another problem is related to the
mode in which the PC power supply
operates. It relies on varying the duty
cycle of the rectified mains pulses applied to the transformer to control the
output voltage. This means that the
secondary rectifier and filter network
must be designed to supply an output
voltage dependent on that duty cycle.
A simple capacitive filtering network
is unsuitable, as it would charge to the
peak secondary voltage regardless of
duty cycle.
The way this dependence is normally introduced is to place an inductor of
appropriate value between the rectifying diode and the capacitor, forming an
LC filter. However, combining an LC
filter with a bridge rectifier does not
clamp the secondary voltage, allowing large spikes to appear across the
primary during transient conditions.
Voltage doubler solution
The schematic diagram in Fig.1
shows a solution to these problems.
It’s based on a voltage doubler circuit
fed by a relatively low secondary voltage, making the secondary winding
easy to fit around the core. A filter
inductor (L2) introduces the duty cycle
dependence necessary for pulse-width
modulation (PWM), while diodes D1
& D2 clamp the secondary voltage,
thereby limiting voltage spikes across
the primary.
There is sufficient space left around
the former for a second 12.6V/2A
secondary to feed the filaments of two
power valves. This winding is also
used to power the switchmode controller circuitry and the cooling fan.
The price we pay for going to the
voltage doubler configuration is reduced power handling. The load current of the centre-tapped configuration
has a large DC component and only
about 20% ripple, whereas in the voltage doubler configuration current must
drop to zero at some point in the cycle.
This means that the average current
is at best half the maximum current.
And since the latter is limited by the
saturation current rating of the transformer, the HV circuit can deliver just
over 60% of the power of the original
supply.
This does not apply to the 12.6V
centre-tapped secondary, however.
So, from an original power rating of
200W, 125W is now available for the
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WARNING!
This is NOT a project for the inexperienced. Do not even think of
opening the case of a PC switchmode power supply (SMPS) unless
you have experience with the design or servicing of such devices or
related high-voltage equipment.
Some of the SMPS circuitry is at full mains potential. In addition,
the high-voltage DC output from this supply could easily kill you.
Beware of any residual charge on the mains and output capacitors,
even if turned off for some time.
The metal case and ground (0V) outputs of all PC power supplies
are connected to mains earth. You should verify that these connections are in place after completing any modifications; under no
circumstances should the output be floated!
DO NOT attempt to modify a SMPS unless you are fully competent
and confident to do so.
HV supply. Alternatively, the unit can
supply 20W for the filament supply
and about 105W for the HV supply.
This is more than sufficient to operate
two power valves.
The circuit is capable of excellent
performance. It maintains full regulation at up to 125W, with ripple at 2V
peak-to-peak, or 0.3% at full power.
This is quite acceptable, as most of
the ripple is at twice the switching
frequency (60kHz) and so is inaudible.
The 100Hz hum component is only
0.08%, which shows the excellent regulation of the TL494, since the rectified
mains source contains 13% of 100Hz
ripple at full power. Over-current protection is retained, with a LED added
to indicate when it is active.
Circuit operation
The schematic of the power section of the HV supply is shown in
Fig.1. The mains input and associated
switching transistor circuitry remain
unchanged, as indicated by the shaded
portion of the circuit.
Typical control circuitry based on
a TL494 PWM controller is shown in
Fig.2. There is quite a bit of variation in
the control circuitry between different
manufacturers, so your circuit might
differ somewhat. This is especially
true if the over-voltage and over-current protection in your supply is based
on the LM339 comparator rather than
on discrete transistors, as shown. Fortunately, there are few modifications
to this part of the circuit.
Operation is quite straightforward,
with the design based on a conventional half-bridge “forward converter”
topology. The 240VAC mains is first
rectified and then filtered by the capacitive divider C6 & C7 to provide
two supplies at ±170V DC. This is
switched alternately through the ferrite transformer by power transistors
Q1 and Q2.
A 1µF capacitor connected in series
with the transformer primary limits
the current by forming an 8Ω load
with inductor L2. This provides some
protection in case of a shorted secondary, which effectively occurs at startup
before C1 & C2 are charged as well as
during fault conditions. Transformer
T3 is used to sense the magnitude of
the primary current for over-current
protection.
The secondary winding develops
a voltage of 502V using the specified
turns ratio. For 400V designs, the secondary voltage reduces to 319V. This
is rectified in the voltage doubler (D3
& D4) and smoothed by an LC filter
(L2, C1 & C2).
During the “on” period, energy coupled to the secondary winding finds a
current path through L2 and into the
load and output filter capacitors. During the “off” period, the energy stored
in L2 is discharged into the load.
The inductance of L2 is chosen so
that current continues to flow for most
of the “off” period at full load. You can
see this effect in the SPICE simulation
(Fig.3). As previously described, the
use of an LC filter ensures that the
output voltage depends on the duty
cycle, as required for PWM control.
Diodes D3 & D4 have to withstand a
reverse voltage of about 900V during
the transistor “on” period, as well as
July 2004 77
The first step is to remove all of the low-voltage components on the secondary
side in preparation for the HV rebuild.
some voltage spikes passed from the
primary to the secondary by interwinding capacitance. Note that these
spikes are generated during the “off”
period by primary leakage inductance they do not transform to the secondary
inductively. Hence, the BYV26G fast
avalanche diode with a peak reverse
voltage of 1400V was chosen for the
job. These are available locally from
RS Components (Cat. 216-9397).
Diodes D1 & D2 provide a low impedance return path for inductor (L2)
current during the switch-off period.
They also combine in the D2-C2-C1-D3
and D4-C2-C1-D1 circuits to clamp the
secondary voltage to ±VOUT.
One of the advantages of this clamping method is that it passes much of
the energy stored in the core of T1 to
the load. This energy would otherwise
recirculate through the primary side
protection diodes (D8 & D9), as well as
dissipate in a more aggressive clamp
or snubber network with higher losses.
At power up, the clamp forms a
short circuit across the secondary until
C1 & C2 are charged, so 100Ω resistors
have been inserted to limit the maximum current. The clamp is important
in reducing the inductive kick of the
primary winding (as opposed to the
primary leakage inductance whose
kick can not be avoided). The effect of
the secondary clamping can be seen as
78 Silicon Chip
a plateau during the “off” period in the
SPICE simulation and the measured
primary voltages of Fig.3 and Fig.4(a),
respectively. This waveform resembles
a square wave at any duty cycle.
An important parameter in the
design of the power sections of the
circuit is the choice of the secondary
voltage to output voltage differential.
This is needed to provide headroom to
compensate for a drop in the secondary
voltage with increased power output,
the difference being made up by the
by duty cycle variation controlled by
the TL494.
Secondary voltage drop has several sources: ohmic losses in inductor
coils, non-linearity of the cores, 100Hz
ripple due to discharge of mains storage capacitors C6 & C7 and voltage
drop across C8 which is charged and
discharged every switching cycle.
The latter two effects contribute to
a primary voltage ripple of 22V and
9V peak-to-peak respectively at full
output power, which manifests as
a 63V ripple across the secondary.
Choosing a 100V differential allows
output voltages of up 800V to be delivered by this supply at full power
and regulation.
The output voltage is smoothed
by a capacitive divider consisting of
two 10µF capacitors (C1 & C2) rated
at 450V. Alternatively, the 400V ver-
sion has a higher output current and
so 47µF capacitors rated at 350V (or
higher) should be used.
At this rating, they are each available in a small package which is easily
accommodated in the space provided
on the PC board for the original 5V
supply components. Their capacitance contributes only about 700mV
of 60kHz ripple (0.1%) at full load.
Two 180kΩ resistors across the
output set a minimum load current,
ensuring that the PWM controller
switches Q1 & Q2 on for at least a small
portion of each period. Resistor chain
R3-R6 divides down the high-voltage
output, developing a lower voltage
feedback signal that is applied to the
non-inverting error amplifier input of
the TL494.
Output voltage regulation is
achieved by varying the duty cycle so
that the voltage applied to the TL494’s
non-inverting amplifier input (pin 1)
equals the voltage on the inverting
input (pin 2). In this case, 2.5V is applied to pin 2 via resistive divider R7
& R8. Hence, if R6=4.7kΩ, then R3 +
R4 + R5 should equal 1311kΩ for 700V
output, or 747kΩ for 400V.
The filament supply is provided
by a simple modified version of the
original 12V secondary. Unfortunately,
we can’t use the TL494 to regulate the
12V supply because the original circuit used a coupled inductor shared
between the secondaries for this purpose. Our two secondaries now have
a high voltage between them. Hence,
an LM350T adjustable 3A regulator is
used to derive the 12.7V supply. It also
powers the 12V cooling fan and must
be fitted with a heatsink.
This secondary also supplies power
to the TL494 via D7 & C5, as in the
original circuit. If a 24V filament
supply is required, the more common
7824 1A regulator can be used, as less
current is required. The cooling fan
can be run from 24V using a 47Ω 5W
series dropping resistor.
When the HV supply is only lightly
loaded, the duty cycle is so small that
the filament supply is not able to deliver its rated current. This can occur
at power-on because no plate current
flows when the filaments are cold.
However, without HV current the filaments can not warm up. To avoid this
stalemate, an auxiliary voltage control
circuit consisting of resistors R13 & R14
and diode D12 is employed.
During normal operation, D12 is
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reverse-biased and the voltage at pin
1 of the TL494 is derived from the HV
supply alone. However, when the filament voltage drops, the cathode of D12
becomes less positive until, at about
1.9V, the diode conducts and prevents
the filament voltage dropping any further. With the resistor values shown,
this threshold is set at about 10.5V.
Circuit protection
Care is required to ensure that the
deadtime control circuit connected to
pin 4 of the TL494 operates correctly
in the modified circuit. The function
of the deadtime control is to provide
over-voltage and over-current protection if the transformer core saturates.
Primary side current is sensed using T3, a small current transformer.
Its primary winding is connected in
series with the primary of the main
switching transformer. T3 employs a
very large transformation ratio (n of
about 180), combined with a relatively
small resistance across its secondary
winding.
This resistance swamps the effects
of primary inductance, such that the
voltage drop across the transformer is
due only to the resistance. The secondary voltage is then proportional to the
primary current at about 2V per amp
with n = 180 and R9 = 350Ω.
The resultant signal is rectified by
D10, smoothed by C10 and applied to
potential divider R10 & R11. When the
voltage at the midpoint of this divider
exceeds about 0.6V, Q3 conducts and a
positive voltage is applied to pin 4 of
the TL494 through diode D11. A voltage of 0V on this pin sets a minimum
deadtime of 4% while at 3.3V, Q1 and
Q2 do not turn on at all.
The values shown for R10 & R11 set
a threshold current of about 2.8A but
you could vary this by altering these
resistors. Transistor Q7 and LED1 were
added to the circuit to indicate activation of the over-current protection.
Transistor Q4 and the voltage divider connected to its base provides
protection against output voltage
imbalances by injecting current into
the base of Q3 under fault conditions.
The voltage divider in the original
circuit was designed to produce
about 0V at the base of Q4 under normal conditions. However, since the
modified supply no longer generates
the negative voltages of the original
circuit but still has the +12V circuit,
this would upset the current balance
siliconchip.com.au
in the resistor network. Catastrophic
failures aside, output voltage regulation prevents over-voltage anyway, so
the easiest solution is to disable this
part of the circuit by removing the associated components (shown shaded
in green on the circuit diagram).
Your circuit may use a different
configuration to the one shown here.
For example, the LM339 comparator
is frequently used for over-voltage
detection. If the voltage on pin 4 of
the TL494 exceeds about 0.3V under
no load, simply disconnect any resistor running from the 12V supply
to the control circuitry connected to
this pin.
Note that some power supplies do
not use discrete components in the
protection circuitry at all. Unfortunately, this article can not hope to
cover all possible variations. If you
do not feel confident in modifying
the existing circuitry, then it is recommended that you construct the circuit
shown in Fig.2 and use it to replace
the protection circuits connected to
pin 4.
This photo and the photo
immediately below show how to
wind and insulate one layer of
the HV secondary. The layer must
start and finish on the secondary
face of the former, adjacent to the
PC board pins. Starting on the pin
end of the former, close-wind one
complete layer (no overlaps). After
the layer is complete, apply about
1 and 1/4 turns of high-voltage
insulation tape. Position the start of
the tape approximately as shown.
Selecting component values
Of major importance in this design is the correct selection of filter
inductor L2. If the inductance of L2
is too small, the circuit reduces to a
standard capacitive voltage doubler
configuration and the dependence of
output voltage on duty cycle is lost.
Alternatively, if it is too large, the
voltage developed across it each halfcycle is insufficient to raise the current
required by the load.
In practice, the optimum value is
about 450µH for a 700V output (150µH
for 400V).
Another challenge is choosing appropriate values for the primary and
secondary damper networks – R2 &
C4 and R1 & C3. The former is needed
to damp the leakage inductance component of the primary, which exists
in all coils due to a small amount of
primary flux that’s not coupled to the
secondary. The energy stored in this
flux during the “on” period (1/2LI2)
generates a current which charges
the transistor output capacitance and
transformer stray capacitance (C0)
when the transistors turn off. In the
absence of resistive losses, this energy
is fully transferred into capacitive
energy (1/2CV2).
If the transformer is rewound as
described in the construction section,
Return the free end of the wire
back to the start side, and then
bind over it with the end of the
insulation tape. The aim is to
insulate the return wire from
the layer beneath and the one to
follow. With the return wire sealed
between the two layers of tape,
continue winding the next layer, or
terminate at the pins if it’s the final
layer.
A view of the completed transformer. Note how the centretapped connection to the final
uly 2004 79
(12V) winding exitsJthrough
a
small hole in the tape, rather than
being terminated at the pins.
D2 & D3
Q1 & Q2
C1 & C2
T1
a large (60W) soldering iron.
Care must be taken when desoldering the ferrite transformer (T1)
to avoid melting the former plastic
and loosening the pins. Remove all
resistors and links leading from the
5V supply to the control circuitry of
the TL494.
L2
Transformer preparation
L1
T3
TL494
REG1
LED1
A view of the reassembled PC board showing the newly rewound transformer
(T1), HV filter inductor (L2) and HV capacitors (C1 & C2). L2 can be secured to
the board using non-acidic silicone sealant.
it will have a leakage inductance of
about 10µH. As this is less than 1% of
the 3.5mH total primary inductance, it
is quite acceptable. C0 is about 270pF,
while at full 150W load, the primary
current can reach 1.7A.
Plugging in the values gives a voltage spike of about 350V. This adds to
the voltage drop across the primary
inductance during the “on” period and
can destroy the output transistors if
the ringing is not damped (protective
diodes D8 & D9 offer limited protection due to their finite resistance and
turn-on time).
The damper network has the side
effect of dissipating energy not only
when the transistors switch off but also
when they turn on. Making the damper
capacitor too large leads to the energy
dissipation at turn-on far outstripping
the parasitic energy.
The parasitic energy is just the energy stored in the leakage inductance
and equates to a power of 0.9W at full
load. We’ve selected a 2W resistor for
R2, which leaves 1.1W to be dissipated
during switch-on. A capacitor of 1nF
will dissipate about 1W in R2 during
the “on” period.
R2 should be 50Ω for critical damping. Making R2 smaller does not
increase damping; rather, the damper
80 Silicon Chip
capacitor effectively acts in parallel
with the primary winding to change
its ringing frequency.
A second source of ringing occurs
when current through L2 drops to zero
during the “off” period. Depending
on the polarity of the half-period, either diode D1 or D2 stop conducting.
However, the voltage across L2 can
not change instantaneously, due to the
energy stored in the diode and switching transistor output capacitance. The
resultant ringing is dissipated by the
damper network across the secondary
and by hysteresis losses in L2.
Construction
You should read the earlier SILICHIP article (October 2003) on
modifying a PC power supply prior to
commencing construction. Note that
quite a few more components need
to be removed here, since most of the
secondary side is unsuitable for HV
operation.
Begin by removing the large lowvoltage secondary capacitors and
inductors. The 5V rectifier and associated heatsink also need to be removed,
as well as the secondary RC damping
network. The multiple power supply
leads for the various output voltages
are best unsoldered and removed using
CON
The next job is to remove the existing windings from the ferrite transformer in preparation for the rewind.
Begin by carefully removing the tape
binding the core sections together, as
it can be reused later. Soak the
transformer in methylene chloride paint stripper overnight to
remove most of sealing varnish.
Note that gloves and protective
eyeglasses should be worn when working with paint stripper.
If you don’t wish to wait overnight,
then the transformer can be warmed
prior to dipping for a few minutes with
a hair drier held at close range. After
about 15 minutes, the transformer can
be gently removed and light pressure
applied by means of a screwdriver
between the slab section of the core
and the former, allowing the latter to
be released.
It is advisable to remove the E-section out of the former immediately by
pressing gently on the centre prong of
the “E” (the outer prongs are fragile and
easily broken). Care needs to be taken
here since this is the only component
that is not easily replaced.
If the E-section won’t separate
with light pressure, then wash the
transformer thoroughly and use a
razor blade and sidecutters to slice
and remove sections of insulation and
copper wire to free it up. Complete the
transformer disassembly by washing
all components and removing all the
wire and insulation from the former.
Transformer rewind
Great care must be taken with the
transformer rewind to ensure primary
to secondary isolation. In particular,
make sure that each layer is completely covered with the tape, right
up to the shoulders of the former, so
that turns from different layers can
not touch.
Except where noted, there should
be no gaps between the start and finish of a layer and the shoulders of the
former. This ensures that wire from the
next layer can not creep into the gap
siliconchip.com.au
and potentially make contact with the
preceding layer.
The HV secondary winding goes on
first, using 28 SWG (0.4mm) enamelled
copper wire. Three layers are required,
producing 117 turns in total. For the
400V version, use three layers of 24
SWG wire instead, producing 75 turns
in total. It does not matter if your winding is a few turns short. The inner layer
is the “hotter” end of the winding. It
must be connected to the third pin
from the edge of the PC board on the
secondary side of the former.
A layer of polyester high-voltage
tape is used to insulate each layer.
Suitable “3M” brand high-voltage
polyester tape is available from Farnell
(cat. no. 753-002). Note that this tape
is 19mm wide, whereas the standard
former requires 17mm tape. To obtain
the correct width, stick strips about
10cm long onto a clean plastic surface (such as transparency film) and
trim off 2mm using a razor blade and
straight edge.
One end of the tape is placed over
the top of a completed layer and the
free end of the wire is returned over the
top and sealed by one turn of the tape
(see photos). The wire must be returned
on the pin face rather than the sides of
the former otherwise there will not be
sufficient room for the core.
The copper strip used in the original
transformer to reduce inter-winding
capacitance is not needed here because
the windings are not interleaved. With
all three layers in place, insulate the
HV secondary with three layers of
polyester high-voltage tape.
Using the same technique, the
primary is now wound in two layers
with 24 SWG wire, for a total of 40
turns. Note that the first layer will be
25 turns, whereas the second layer
will be only 15 turns. This leaves a
gap between the finish of the winding
and the shoulder of the former. Before
applying the inter-winding insulation
over the second layer, this gap must be
filled in with tape.
To achieve this, cut strips of highvoltage tape of the appropriate width
and build up the gap to the same height
as the windings. The idea here is to
achieve a smooth, level surface for
the final winding. That done, insulate
the primary with two layers of highvoltage tape.
Finally, the 12V secondary is wound
with 12 centre-tapped turns in a single
layer using 24 SWG wire and insulated
siliconchip.com.au
Fig.3: the output from a SPICE simulation of transformer primary voltage and
toroid current waveforms. The simulation results closely follow the actual
waveforms measured in the working prototype.
with a single layer of high-voltage
tape. It’s easier if the centre-tap connection is not terminated at the pins
(see photo).
The transformer core can now be
fitted, making sure that the abutting
faces are perfectly clean. This is necessary because the ferrite core is of
very high permeability material (ie,
µe about 2000). An air gap of only one
two thousandths of the core length
(about 25 microns) will be sufficient to
halve the coil’s inductance. The core
sections are pressed together tightly,
bound with the original tape, and the
whole assembly sprayed with lacquer
and left a few hours to dry.
It is best if the former pins are
masked with tape prior to spraying to
make subsequent soldering easier.
Toroid rewind
Toroid L2 is wound next. A key
requirements for L2 are that its insulation should withstand about 500V and
it must be able to dissipate the heat
generated by hysteresis in its core.
The latter is not to be confused with
ohmic losses in the windings (which
are small here) and arises because
the core does not demagnetise at zero
current. This remnant magnetism is
removed by reverse current every cycle
and manifests itself as heat. In practice, hysteresis losses can be reduced
by using a larger core size for a given
value of inductance.
With this in mind, L2 consists of two
standard 25 x 10mm toroids glued at
the faces to form a single core. This
gives the required inductance in a
single layer and reduces hysteresis
heating. Suitable core material is the
standard yellow/white or green/yellow ferrite typically used in PC power
supplies.
The original low-voltage windings
are discarded and the faces of the
toroids thoroughly cleaned before gluing. L2 is wound in a single layer with
56 turns of 24 SWG wire. For 400V
versions, use 33 turns of 24 SWG wire.
In operation, the core should only
get warm to the touch at full power
(make sure you turn off the converter
before checking this!).
PC board rebuild
All of the necessary components can
now be installed on the PC board.
The HV section occupies the area
previously taken up by the 5V supply.
Although the exact PC board layout
varies between manufacturers, a typical design allows easy accommodation
of all the components shown in Fig.1.
However, you might need to break a
few copper tracks with a sharp knife
or engraver tool and add a few links
with insulated hook-up wire.
Important: be sure to leave at least
1mm clearance between all high-voltage tracks in this part of the circuit.
Remember to install the resistive
divider from the HV supply to pin 1
of the TL494 in place of the original
divider running from the 5V supply.
Inductor L4 and capacitor C9 are
mounted directly across the rear of
the output terminals (see photos).
July 2004 81
Fig.4(a): this scope waveform was measured across
the primary of transformer T1 and shows the alternate
switching of transistors Q1 and Q2. Notice how the
secondary voltage clamp has flattened the peaks
of the waveform to produce a square-wave voltage
that’s independent of the duty cycle. Note also that
the waveform peaks are slanted slightly due to the
discharging of C8.
Fig.4(b): the voltage across toroid L2 over several cycles.
The peaks of about 370V occur during the “off” period
when L2 discharges into the smoothing capacitors (C1
& C2). Some ringing occurs when the current drops to
zero, as described in the text. During the “on” period,
the voltage across L2 equals the difference between the
secondary and output voltages, decreasing steadily as
C8 is charged.
Fig.4(c): 60kHz output ripple at full load is about
2V p-p at 700V DC.
Fig.4(d): 100Hz hum can be seen on top of the 60kHz
ripple and amounts to only about 0.6Vp-p.
Note: for safety reasons, these waveforms were all taken with the SMPS connected to the mains via an
isolation transformer. Don’t attempt this unless you know exactly what you are doing.
The inductor consists of 5-6 turns of
hookup wire wound around a small
toroid.
The 12V circuit occupies its previous location but make sure that all
components not shown in Fig.1 are
removed. The 4µH inductor (L3) can
be salvaged from the original 5V supply and consists of 7-8 turns of copper
wire around a ferrite rod.
Once you are certain that no HV
is fed anywhere except as shown in
Fig.1, you are ready to apply power to
the circuit. It would be useful to have
82 Silicon Chip
a load available to check operation at
reasonable power levels. I used several
strips of five 4.7kΩ 5W resistors connected in series to provide a 25W per
strip load at 700V.
Warning: switchmode power supplies have been known to explode on
failure, expelling particles of component material such as metal, epoxy
and glass at high speed. Close the case
or wear protective eyeglasses before
applying power!
If the circuit fails to deliver substantial power, the problem might be
due to the current protection circuit.
Check that the voltage on pin 4 of the
TL494 does not exceed about 0.3V under normal load. If it does, this part of
the circuit is malfunctioning. Follow
the techniques described in the circuit
protection section above to track down
the problem.
Finally, because the modified converter is less efficient than the original,
it requires better cooling when operating at full power. This can be achieved
by switching the cooling fan around so
that it forces the air into the case. SC
siliconchip.com.au
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