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How to use the TEA1100 fast nicad charger IC The TEA1100 nicad charger IC, as used in our recent Fast Nicad Charger project, has a number of interesting features which put it out in front. These include digital voltage sampling and filtering as well as switchmode or linear operation. We look at these in detail and go through some design examples. By DARREN YATES & LEO SIMPSON The Philips TEA1100 Battery charger IC is a 16-pin DIP package which contains everything to produce a simple yet highly integrated battery charger for nickel cadmium (NiCd) and the new nickel metal hydride (NiMH) batteries which have a higher capaci­ty than nicads. The TEA1100 has three methods of guarding against over-charging: temperature detection, clock timeout and an advanced form of voltage detection referred to as “dV sensing”. Supply voltage Unfortunately, the chip has a fairly awkward supply voltage range, which is between 5.65V and 11.5VDC. This makes it not quite suitable for car operation without supply regulation cir­cuitry and not low enough to permit operation from a 5VDC regula­tor. However, a 7808 or 7809 regulator will be more than adequate and, in fact, you can even get away with just a standard zener diode/transistor buffer voltage stabiliser such as that used in the Fast Nicad Charger design published in the May 1994 issue of SILICON CHIP. Linear or switchmode As we mentioned in the introduction, the TEA1100 IC can run in linear or switchmode operation. The benefit of the switchmode option is the efficiency which can be gained by charging lower voltage cells from a higher Fig.1: this circuit using the TEA1100 in switchmode was the basis for the Fast Nicad Charger published in the May 1994 issue. The design example in the text will enable you to tailor the circuit to your application. 6  Silicon Chip Fig.2: this linear version of the TEA1100 circuit could be used in RF-sensitive applications. In this case, the output of the chip is taken from pin 2 rather than the PWM output of pin 1. voltage supply rail. This feature was used in our Fast Nicad Charger project. If you’re wanting to charge Nicads in a noise sensitive application then you can easily set the IC up to charge in linear mode, greatly reducing the circuit noise. The linear circuit uses less components than the switchmode circuit but has considerably more heat dissipation, as you would expect. Fig.1 shows a sample switchmode circuit, very similar to that featured in our May 1994 issue. Fig.2 shows a linear charge circuit, powered from the 240VAC mains supply. And finally, to give some idea of the chip complexity, Fig.3 shows the block diagram of the TEA1100. dV sensing Instead of comparing the voltage of the battery being charged to a static voltage reference, the TEA1100 uses a dynamic process called “dV sensing”. The “dV” term comes from calculus and refers to the process of looking for a very small change in battery voltage. The TEA1100 compares the present battery voltage to the previous sampled voltage and checks for a 1% drop. The theory behind this is that when a nicad is being charged, its voltage rises very gradually towards full capacity but once past this point, the battery voltage begins to drop slightly. If a battery charger circuit does not look for this voltage drop, it will never give an optimum charge – it will either under or over-charge. Either way, the battery life will ultimately be reduced. Fig.4 shows the characteristic rise in battery voltage during charge and the slight droop as it reaches full charge. The TEA1100 ends the charge cycle upon sensing a 1% drop in the battery voltage. That amounts to about 16mV for a typical nicad cell. Now you might be wondering how they manage to reli­ably detect 16mV when the circuit lines could be subject to all sorts of noise and switching frequencies, if the chip is being operated in switchmode. The answer lies in the method of sampling the battery voltage. The PWM (pulse width modulation) is disabled for 10 clock cycles, after which the sample and hold amplifier takes a meas­ urement of the battery voltage. This way, the noise generated by a “ringing” or decaying supply rail is removed and a much greater degree of accuracy maintained. The 10-cycle delay gives sufficient time for the inductor to stop ringing but it does mean that the inductance must lie within a particular range – it must be high enough in value so that it will perform its job as an inductor in a switchmode circuit but it must be small enough in value so that the supply rail is quite stable by the time 10 clock periods have passed. We’ll talk about this more a little later. The dV sensing comes under the block entitled “battery full detection” in the diagram of Fig.3. As already noted, the TEA1100 does not compare the battery voltage to a static reference. Because it is a dynamic process, the monitored input voltage need only be between 0.385V and 3.85V. This is fed to pin 7 which is labelled “VAC” for Voltage ACcumu­lator. The way it works is like this: The VAC voltage is sampled at a rate equal to the clock frequency divided by 216. Each VAC voltage sample is digitised and stored in a register with a quoted resolution of 12.5 bits. At the time of the next sample, the stored value is converted back to an analog voltage and compared with the voltage on the VAC pin. If the VAC voltage is higher than the stored value, then this new value is digitised and stored in the register, overwrit­ing the previous value. If not, the previous value remains in the register. The circuit then checks for a 1% drop as we mentioned before and if found, switches the circuit to trickle mode and flashes the LED (connected to pin 15) to indicate that the bat­teries are fully charged. This clever mix of analog and digital circuitry results in a dynamic process which takes the battery’s physical characteris­ tics into consideration. Since no two nicads charge up to exactly the same voltage, this relative method provides accurate “full” detection for all cells, regardless of their final voltage. Incidentally, much the August 1994  7 VP 12 Vref 10 VS 6 NTC 3 Rn 11 IB 5 CP 9 Vr1 SUPPLY GND 16 Vhigh PROTECTION Vr3 MAINS ON RESET V In LSP Iref > t AO 2 A2 Vr2 Vlow PROTECTION Vr4 LS 4 A1 > > OSC DISABLE TIME OUT > R s+h BATTERY FULL DETECTION VAC 7 PWM 1 PWM & R TIME OUT PROTECT > LED 15 R 1/10 OSC TO PWM :1:2:4 PRESCALER COUNTER CONTROL CURRENTLESS SENSING AUX PULSES 13 OSC 8 PR 14 SYNC Fig.3: the block diagram of the TEA1100. This complex chip senses the small drop in voltage which occurs at the end of charge for nicad & NiMH batteries, so that the charger can be automatically switched off. same monitoring method was used in the “Fast Charger for Nicad Batteries” featured in the January and February 1991 issues of SILICON CHIP. The beauty of the dV sensing system is that the VAC input (pin 7) can be anywhere between +0.385V and +3.85V. This means that the VAC resistor divider network can be the same whether you wish to charge two or 10 cells, or any number of cells in between. To satisfy this condition in the circuit of Fig.1, R14 should be 47kΩ while R15 should be 10kΩ. C8, the input filter capacitor, can be 10µF 16VW. Note that to satisfactorily charge 10 cells, you will need an input voltage of at least 22V DC when in switchmode because the maximum pulse duty cycle is 78%. The above is based on an overvoltage level of 1.7V/cell and a nominal battery voltage of 1.2V/cell. The VAC input has four voltage thresholds which determine the chip’s behaviour. Firstly, below 0.3V, the IC assumes a short circuit (crook) battery and switches to trickle charge mode; above 0.385V and below 3.85V, the IC uses the dV voltage detec­tion method 8  Silicon Chip to determine the charge state; and finally, above 4.25V, the IC assumes open circuit or no batteries present and switches off. The impedance of this input is greater than 200MΩ. Note too that for charging just one or two cells, the VAC input (pin 7) can be connected directly to the cell(s). Output voltage This brings us to an important feature of the Fast Nicad Charger published in May 1994 and one which has caused confusion to many constructors of this circuit. Since the circuit relies on dV sensing to end the fast charging mode, it goes without saying that it will not work unless it is actually charging cells. If you attempt to test the circuit without a nicad battery load, it will switch off. Our testing instructions for the above circuit would have added to this confusion by referring to an open circuit output voltage test. The point is that you cannot test the charger’s output voltage unless cells are connected. If you attempt to simulate the presence of cells with a large electrolytic capaci­ tor, the output voltage will rise until pin 7 reaches +4.25V whereupon the circuit will switch off. In fact, the circuit of May 1994 does not even need the switch to select between two and four cells. The switch setting for two cells can be omitted and then circuit will happily charge two, three or four cells in series without further modifications. In admitting this mistake, we can only plead that it only become obvious after close reading of the copious application information which Philips has made available on the TEA1100. 0.5% detection In some cases, such as “fast-charge” nicads and NiMH cells, a dV of 0.5% is more appropriate due to the higher level of input charge current they can tolerate. This IC can provide charge rates up to an incredible five times the battery capacity or “5C”. An example of this would be charging a racing pack in about 15 minutes. This increased sensitivity can be easily achieved by in­serting a zener diode of about half the battery voltage into the sensing resistor string. An example of this can be seen in Fig.5. The zener diode is selected to be about half of the fully charged battery voltage, based on a level of 1.7V/cell. Protection Apart from the active protection features already men­tioned, the TEA1100 features under-voltage shutdown and tempera­ture sensing with a thermistor input circuit. The first of these, the under-voltage shutdown, activates when the supply voltage falls below 5.25V. In this case, the IC goes into a “power down” mode in which it becomes non-active and draws around 35µA (45µA maximum). The second form of protection involves a negative tempera­ture coefficient (NTC) thermistor to monitor the temperature of the battery during charging. This feature wasn’t included in our May 1994 project to keep the construction simple. In practice, where this feature is used, the therm­ istor is incorporated into the battery pack and is automatically connected when the battery is put on charge. The temperature monitoring feature is recommend­ed for batteries which need to be recharged as soon as they have been removed from their load. The classic example of this is 1200mA.h racing packs for electric model aircraft and cars. The drain on these batteries is very high - often tens of amps or more – and so they will be quite hot (or even stinking hot!) when they are removed from the load. The danger is that if you fast-charge a hot nicad battery, you can damage it. The temperature protection provided by the TEA1100 prevents fast charging from occurring while the battery temperature is outside the specified range. The NTC thermistor is featured on the circuit of Fig.1 and is connected to pin 3. If the thermistor is not required, it can be omitted from the cir­cuit, together with R11. Fig.4: the voltage characteristic of a 2-cell nicad battery back during charge. If charging continues beyond the droop in voltage, cell damage can occur. VOLTAGE (V) For example, for a 6 cell pack, the maximum voltage is 6 x 1.7V = 10.2V, so a zener diode of 5.1V would be suitable. The maximum voltage level the VAC input will now see is 5.1V, so the input resistor divider must now be recalculated accordingly. R14 on Fig.1 could then be reduced to 22kΩ. CHARGE TIME (MINS) capacitor connected to pin 13. This timeout period is usually set to about 125% to 150% of the expected fast charge time but in critical high charge rate applications, you can set it to the expected charge time (100%). In practice, the timeout period should only be set by adjusting the capacitor (C at pin 13), as varying the reference resistor will change other circuit parameters. Design example The easiest way to understand how to use this IC is to go through a design example, using the circuit of Fig.1. This way, you’ll get an idea of what has to be done and the order in which you have to do it. Let’s say we wanted to design the timeout circuit to run a charger which will charge up a set of four nicad cells in one hour. If we use the 150% rule, then our timeout period, tTO, will be Timeout counter Finally, there is the backup protection of a timeout coun­ter, which automatically shuts down the charger after a time equal to 226 times the clock period, has expired. The clock period is determined by the reference resistor connected to pin 10 and the timing Fig.5: a zener diode equal to half the fully charged battery voltage can be added to the circuit to enhance the dV sensing capability so that it will detect a drop of 0.5%. 1.5 x 60 mins = 90 mins. The timeout period is determined by the following formula: tTO = 226 x Tosc x p where Tosc is the clock period and p is a prescaling factor which you can program to be either 1, 2 or 4, depending on how you connect pin 8. By leaving pin 8 open, you set the prescaling factor to 2. Connecting it to pin 6 sets it to 1 and pulling pin 8 to ground sets it at 4. The beauty of this system is that it allows you to have three different charge periods without having to change the timing components. For our example, let’s connect pin 8 to pin 6 to set the prescaling factor (p) to 1. The oscillator frequency (1/ Tosc) now needs to be 12.4kHz (ie, Tosc = (90 x 60)seconds/226). As mentioned be­ fore, this frequency is set by the time constant formed by the reference resistor Rref (R13) and the oscillator capacitor Cosc (C7) based on the following equation: Tosc = 0.93(Rref x Cosc) Now Rref is chosen to be within the range of 12.5kΩ and 125kΩ based on the necessary charge current. In our example, let’s assume that the resistor is 27kΩ. Plugging this value into the above equation gives a value for Cosc of .0032µF which we can quite happily round to .0033µF. Charge current settings OK. Let’s say that we wish to charge our batteries at a fast rate of 700mA. R4 and R8 are used to set the current. R4 should be a 5W type. You have some leeway in picking the value of this resistor, so long as its value August 1994  9 Using the TEA1100 fast nicad charger IC results in a voltage drop of between 50mV and 200mV when the circuit is in fast charge mode. You can work out a suitable value for R4 from the following equation: Vcs = Ifast x Rcs where Ifast is the fast charge current and Rcs is R4. In our design example, 0.1Ω will give us 70mV which is within the de­sired range. R8 is referred to as the fast charge current set resistor Rfc and it can be calculated from the following equa­tion: Rfc = (Ifc x Rref x Rcs)/1.25 where Ifc is the fast charge current rate, Rref is the 27kΩ reference resistor R13, and Rcs is the 0.1Ω current sensing resistor R4. By using this equation, we get a value for Rfc of 1.512kΩ, so a 1.5kΩ 1% resistor will be perfect for R8. determined by the worst case ripple current at the trickle current setting and follows this equation: Lmin = Vo’max(1-delta)Tosc/2Iav where Vo’max is the maximum battery voltage plus the forward diode voltage drop. For four cells, this works out to be 6.8V + 0.7V = 7.6V. This is based on the fact, that the maximum voltage per cell will be 1.7V; “delta” refers to a charge current duty cycle of 50%. So, using the above equation, we get a minimum inductance value of: Lmin = 7.6 x (1-0.5) x 80 x 10-6/2 x 0.35 = 434µH. Hence the inductor can be anywhere between 5mH and 434µH. Why not go for the perfect compromise and settle upon 2mH? What inductor? Winding an inductor presents many constructors with a problem since they don’t have access to the necessary information involving readily available toroids. Indeed, a comprehensive article on this subject alone could take many pages. However, to keep it simple, we’ll just deal with the three readily available iron powder toroids made by Neosid and available from Altronic Distributors and Jaycar Electronics. The general formula for inductance using these toroids is: n = 1000 √(L/AL) where n is the number of turns, L is the inductance in millihen­ries (mH) and AL is the inductance factor of the particular core. For the smallest core, Neosid 17-732-22, 14.8mm OD, AL is 44; for the medium core, Neosid 17742-22, 33mm OD, AL is 59; and for the largest core, Neosid 17-745-22, 44mm OD, AL is 116. Having calculated the number of turns to obtain the re­quired inductance on the core of your choice, you then must check whether it is likely to be saturated at your proposed operating current. To do this, we calculate the core energy with the fol­lowing formula: E = LI2 where E is measured in joules, L is the inductance in henries and I is the current in amps. For the three cores Earlier on, we mentioned that with switchmode operation, you have to be careful in selecting the value of the inductor – too low a value will result in the circuit not working efficient­ly and too high a value will result in the dV sensing circuitry picking up remnants of the switching voltage due to the “ringing” effect of the inductor. For this dV sensing to work, the induc­ tance current should have decayed to zero within nine clock cycles. So the maximum inductance is set by the following equa­tion: Lmax = 9 x Tosc x (Vo + Vf)/Io where Tosc is the period of the clock frequency, Vo = the flat battery voltage (around 1V per cell) plus the voltage drop across the fast recovery diode D2 (usually taken as 0.8V) plus the voltage across the current sensing resistor. Io is the average current through the inductor which is a fast charge current. In the example we’ve been working through, this would give us a maximum inductance of: Lmax = (9 x 80µs x 4.8V)/700mA = 5mH. This assumes four cells with a flat voltage of 1V each, plus the 0.8V drop for the fast recovery diode, D2. The 80µs figure is the clock period at 12.4kHz. The minimum inductance value is 10  Silicon Chip Winding an inductor under discussion, the maximum stored energy levels are 0.71mJ for the 14.8mm OD core; 5.1mJ for the 33mm OD core; and 16mJ for the 44mm OD core (OD stands for outside diameter). If the core you have chosen will saturate at the required current and inductance, then you will have to use a bigger core. One final point must be covered here before we leave the subject of induct­ors and that is that the actual current flowing in the induc­tor referred to in the formulas above is the pulse current; it is not the charging current. Typically, the pulse current will be twice the average charging current. Trickle charge When in trickle charge mode, the TEA1100 continues to pulse the battery with the fast charge current but at a much lower duty cycle. As it seems with just about everything else on this IC, you have a choice of one of two ways to set the trickle current, depending on how you connect pin 11, designated the “Rn” input. The first method is to leave pin 11 unconnected. In this case, the repetition and duration of the trickle current pulses is determined by the chip itself. The repetition rate is set as 2-14 x tTO = 330ms in our example. The duration time is set to 0.75 x 29 x Tosc, where Tosc is the clock period. In our example, this works out to be 31ms. This also gives us a duty cycle for the trickle current of 9.4%. The average trickle charge current based on this duty cycle is set by the following equation: Itrickle = Ifc/2 x duty cycle = 30mA. The second method is to set the average trickle current yourself by connecting a resistor Rn to pin 11. The rule for this resistor is that it must be within the range of 25kΩ to 250kΩ and must be greater than the reference resistor Rref. The new trickle current equation looks like this: Itrickle = Ifc x (Rref/Rn) x duty cycle With Rn equal to Rref (27kΩ), the trickle current is 60mA and 7mA with Rn equal to 250kΩ. Linear design example Let’s say that we want to charge three “AA” cells in one hour, using the circuit of Fig.2. The required TABLE 1 Number of cells to be charged Transformer secondary voltage (V RMS, full load) Capacitor value (µF/A) Capacitor voltage rating (VDC) 2 7 4000 16 3 9 3000 25 4 11 2400 25 5 13 2000 35 6 15 1700 35 7 17 1500 40 8 19 1300 40 9 21 1200 50 10 23 1100 50 current is based on the following equation: Iout = (A.h x 60 x 1.4)/charge time (mins) So for a 600mA.h battery, the current would need to be: Iout = (600mA.h x 60 x 1.4)/60 = 840mA In case you’re wondering why it just isn’t 600mA, the reason is that there are substantial losses in the battery when charging takes place, so you need to increase the charge current by 40% to make up for these losses (ie, heat etc.) At this current, the main pass diode D5 can still be a 1N4004 but the transistor will have to be something like a TIP32C, a device which can handle the current and the power dissipation. And it will need a heatsink. Power dissipation Table 1 gives the required transformer secondary voltage and the suggested capacitance per amp of required current and voltage rating of the filter capacitor. Now for our design exam­ple, to charge up three cells, we need a transformer secondary vol­tage of 9.1V. The power dissipation can be found from the follow­ing equation: Pdiss = 1.3 x Iout x (Vsec - 2.0) = 1.3 x 0.84A x (9.1 - 2.0) = 7.8W Basing this on a maximum temperature rise of 55°C above ambient, the required heatsink will have to be better than 55°C/7.8W or 7°C/W. Now obviously, this is quite a bit of power being wasted so you will have to decide whether the need for a linear charger outweighs the benefits of the switchmode alternative. OK, so we’ve determined the cur- rent we require and now we have to tell the TEA1100 what we want. To do this, we again start with a reference resistor of 27kΩ, just as for the switchmode version. Next, we have to choose the main current sensing resis­tor (R1) and again, for our charge current of 840mA, a 0.1Ω 5W resistor will give us 84mV which is good enough. Remember that this resistor doesn’t set the current on its own. This is done by resistor R3 on the circuit. This resistor is determined by the following equation: Rfc = (Rref x Rcs x Ifc)/1.25 and in our design example, R3 becomes: R3 = (27kΩ x 0.1Ω x 0.84A)/1.25 = 1.814kΩ A value of 1.8kΩ will be close enough. Trickle charge As with the switchmode version, the trickle charge current can be set to just about anything you want. By connecting the prescaling pin (pin 8) to pin 6 and leaving resistor R7 open circuit, the TEA1100 will automatically set the trickle charge current to 1/20th of the fast charge rate. In our example, this would work out to be 42mA. Now this may be too high, in which case, you can change the trickle current by connecting resistor R7 from pin 11 to ground. The relationship between this resistor and the trickle charge current is set by the following equation: R7 = (1.25 x Rfc x 0.094)/(Itrickle x Rcs x p) Let’s say we wanted the trickle current to be 15mA instead of 42mA. By working through the above equation, resistor R7 would need to be: R7 = (1.25 x 1.8kΩ x 0.094)/(15mA x 0.1Ω x 4) = 35.2kΩ. A 36kΩ 1% resistor will get you fairly close to the mark. You should note a couple of things here. Firstly, we’ve had to change the prescaling factor to four. Now the reason for this is that the prescaling factor not only works on the timing circuitry but also on the charge current ratio; that is, the ratio of the fast charge current to the trickle charge current. With a prescaling factor of one (pin 8 to pin 6), the maximum ratio is 20:1. For a prescaling factor of two (pin 8 open circuit), it is 40:1 and for four (pin 8 to ground), it’s 80:1. Now for our design we want a ratio of 840mA/15mA = 56:1. Setting the prescale to either one or two won’t get us this value so we have to go to a prescale factor of four. The reason for the change is that if resistor R7 is greater than twice the reference resistor R6, then the IC automatically selects half of the fast charge reference current. This gives us our maximum 20:1 with a prescale of one, 40:1 with p set to two and 80:1 with p set to four. In most situations, resistor R7 should not be less than the reference resistor. If by working through the equations, you find that R7 is less than R6, either change the prescaling factor or remove the resistor from the circuit altogether. Timeout counter settings The last thing to do is to set the timeout period and since we have already set the reference resistor R6 to 27kΩ, the only component value which affects the time is capacitor C4 and this can be determined by the following equation: C4 = (60 x timeout)/0.93 x Rref x p x 226 Getting back to our design example, let’s say that we’re happy with a trickle current of 42mA and we want the timeout period to be 60 minutes. Capacitor C4 then works out to be: C4 = (60 x 60)/(0.93 x 27kΩ x 1 x 226) = .00213µF (.0022µF will be close enough). Note too that this capacitor value will change if you change the pre­ scaling factor as in the above example where we looked at a trickle current SC of 15mA. August 1994  11