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In this chapter, we will deal with oscilloscopes using monoacceleration tubes and up to 20MHz bandwidth. High voltage circuits, DC coupled blanking/ unblanking and triggering methods are investigated in some detail. By BRYAN MAHER If you are puzzled by some strange fault in any electronic equipment and your voltmeter gives no clear evidence, your first question should be “what does the oscilloscope show?” It can reveal at a glance more information than all the voltmeters in the world can demonstrate. Maybe you have subtle supersonic oscillations. To see some faults, even in audio equipment, your CRO may need a bandwidth of 20MHz or more but whatever the band­width, a CRO is a very handy instrument. Last month, we saw the basic configuration of a cathode ray tube (CRT), as shown in Fig.1. The heated cathode emits electrons which are attracted forward by the (relatively) positive poten­tial on the acceleration grid (G3) and the conductive aquadag coating inside the tube, near the screen. When these fast electrons hit the fluorescent phosphor coating on the inside of the front glass screen, light is emitted. The resulting trace on the screen is a graph of the voltage signal we apply to vertical deflection plates Y1 and Y2 via the vertical amplifier. The electron beam current is determined by the tube, its acceleration voltage and your setting of brightness; typically between 10 and several hundred microamperes. In the simplest arrangement, as in Fig.1, after hitting the screen, the electrons must leak across the phosphor to the conducting aquadag and then to ground. G3 is called the acceleration grid. In this simple tube, it has the highest positive potential. The word grid is used here (even though it is posi- This photo shows the base ends of two elementary CRO tubes with the glass envelope removed. In each, nearest the base is the electron gun, consisting of heater, cathode, control grid G1 and hollow tubes we call focus grid G2 and acceleration grid G3. Further from the base, ceramic insulator pillars separate and support the pair of vertical deflection plates. Farthest out are the two horizontal deflection plates. 56  Silicon Chip Fig.1: this sort of CRO tube is a monoaccelera­tion type because all electron beam acceleration occurs before deflection. Therefore this type of tube requires high voltage signals of up to 250 volts swing applied to the deflection plates. tive) because electrons pass straight through it. The term anode is reserved for electrodes which collect electrons. The CRO tube shown in Fig.1 is known as a monoacceleration type, because all acceleration of the electrons is achieved before beam deflection occurs. We will see how this fact limits the realisable bandwidth to about 20MHz and acceleration voltages to the 2kV to 5kV range. In Fig.1, to prevent deceleration of the electron stream, G3, the deflection plates and the screen are all maintained at about the same potential. But the deflection plates are low voltage circuits. Therefore, we choose to ground the high voltage supply at the G3-screen end; ie, its positive side. The heater, cathode K, control grid G1, focus grid G2 and acceleration grid G3 are collectively known as the electron gun. Because the high voltage supply in Fig.1 is positive grounded, the cathode K is at a high negative potential with respect to earth. But an even greater negative potential is applied to control grid G1. This negative bias (ie, the K-G1 potential difference) determines the beam current and thereby varies the brightness of the trace on the screen. The high voltage supply usually consists of a high frequen­cy oscillator driving a ferrite core step-up transformer, fol­lowed by high voltage rectifier(s) and filter capacitors. High frequencies are chosen for four reasons: (1) any sounds from the transformer core are supersonic, above human hearing; (2) a high volts-per-turn ratio is easily achieved; (3) the trans­former can be small and light; and (4) only small filter capaci­tors are required to smooth the rectified current to DC. Deflection options An electron beam can be deflected by an electrostatic field between two deflection plates or by a magnetic field at right angles to the path of the electron beam. Almost all analog oscilloscopes use electrostatic deflection, as in Fig.1. There are two reasons for this: (1) deflection of the electron beam is linearly proportional to the voltage applied to the deflection plates; and (2) the low capacitance between the vertical deflec­tion plates (about 2pF) can be easily driven over a very wide range of frequencies, from DC to 1200MHz (1.2GHz) or even higher, assuming suitable amplifiers. By contrast, magnetic deflection requires large signal currents flowing in coils (the yoke) wrapped around the neck of the CRO tube. This is unsuitable for analog oscilloscopes for the main reason that the inductance of the yoke windings severely limits the current as the frequency rises. Magnetic deflection is universally used in TV and computer monitor CRTs but here the deflection frequencies are fortunately quite low and fixed: 50Hz vertical and 15625Hz horizontal, in the case of PAL TV. This allows each deflection circuit to be optimised for its particular frequency. Electrostatic deflection For parallel deflection plates, the distance across the CRO screen (vertically or horizontally) that the electron beam is deflected is directly proportional to: (1) the potential difference Vd between deflection plates; (2) the dis­tance Ls from the deflection plates to the screen; and (3) the length Lp of the deflection plates. In addition, April 1996  57 Fig.2: simplified diagram of the high voltage circuits suitable for a small analog oscilloscope. Transformer T1, operating at 60kHz, provides two independent negative DC supplies. The -1.5kV supply at TP3 provides the electron beam current from cathode K to screen. The -1.6kV supply at TP2 is dedicated to providing the control grid G1 potential. it is inversely proportional to the accelerating voltage VHT between the cathode and the deflection plates and the spacing “d” between them. These factors come together in the following equation for Deflection Factor which gives the deflection voltage required for one centimetre of trace length on screen: Deflection Factor = Vd/cm = (2d.VHT)/ (Ls.Ld) volts/cm This equation dictates that the vertical deflection plates should be placed as far from the screen as possible. Why? To correctly display the signals, the frequency response of the vertical system needs be much higher (often 20 times more) than the horizontal. Therefore, the design of the vertical ampli­fiers is much more critical, in terms of bandwidth, than the horizontal amplifiers. And it is easier to obtain high frequency response from any amplifier if less output voltage is required. From the equation we see that, for a given length of trace across the screen, less voltage is required at the deflection plates farthest from the screen. Therefore, the vertical plates are always furthest from the screen. Of course that means more deflection voltage is needed at the horizontal 58  Silicon Chip deflection plates as their distance to the screen is less. This is usually not a problem, due to the lower band­width demanded of the horizontal sweep system. The above equation also indicates that by lengthening the vertical deflection plates, we could achieve deflection with less output voltage from the vertical amplifier. That certainly is practised but cannot be overdone because longer plates mean greater inter-plate capacitance which must be driven by the vertical amplifier without loss of frequency response. Furthermore, long plates mean that at high enough frequen­cies the signal will cycle to the opposite phase while any one elec­tron is still between the plates, partly cancelling the deflec­tion achieved and increasing the plate current. In modern CRO tubes, the vertical deflection plates are commonly long and curved, as a compromise between these conflicting factors. For a really bright, sharp trace on the screen, high accelera­tion voltages must be used but the above equation says that higher VHT results in smaller deflection angles. This is because faster electrons are more difficult to deflect. Typical deflec­tion angles for CRO tubes are only 10-30°. Because of this, typical CRO tubes tend to be much longer than their diamet­er. Diameters commonly range from 50-135mm, with lengths from 200-600mm. Magnetic shielding In all equipment using CRO tubes, the power transformer should be carefully positioned to avoid accidental deflection of the beam by 50Hz magnetic fields. As well, electrostatic CRO tubes are usually shrouded in a shield of mu-metal, to prevent interference to the electron beam by stray magnetic fields. TABLE 1 Acceleration Pot. Electron Velocity 2kV 26,400km/s 5kV 41,600km/s 10kV 58,400km/s 20kV 81,500km/s 75kV 147,000km/s 120kV 176,000km/s Electron speeds Electrons accelerate all the way from the cathode to the region of highest positive potential. In monoacceleration tubes, this means electrons continuously gaining velocity between K and G3. They then coast at constant speed to the front screen. Great­er velocity Fig.3: timing diagram for the CRO tube horizontal deflection and trace brightness control. Sections of the repetitive input sinew­ave signal actually displayed on screen during the forward sweep are from t1 to t3, t11 to t13 and so on. During the remainder of time the screen is blanked to conceal the retrace and holdoff and wait times. results from using a higher accelerating voltage. Table 1 shows some examples. Deflection factor The design of any analog oscilloscope must start with the vertical deflection factor of the tube; ie, the number of volts that must be applied between the deflection plates to produce one centimetre of trace on screen. The lower this value, the easier is the design of the vertical amplifier and the wider the bandwidth that can be achieved. One of the earliest CRO tubes, famous in Australian Radar sets during World War 2, was the ubiquitous 5BP1 (125mm in dia­meter). Cheap in postwar disposals stores, this tube found its way into many home constructors’ projects. It had the disadvantage of a high deflection factor value. With 2.2kV acceleration voltage, the 5BP1 required a 320V peak-to-peak signal between the vertical deflection plates to draw a line 8cm high; a vertical deflection factor of 40V/cm. If the acceleration potential on similar tubes was raised to 5kV to produce a brighter trace on the screen, then a deflec­tion voltage swing of about 700V would be required to pro­duce an 8cm trace; ie, 88V/cm. A deflection amplifier capable of producing such a large output voltage swing, even at only 2MHz bandwith, would be very difficult to design. Later tubes progressively reduced this demand for high deflection voltages. The European types 30C3 and 30E7, with 4kV acceleration potential, had a deflection factor of 50V/cm. Today, to keep the deflection factor low, monoacceleration CRO tubes are sometimes limited to a high voltage of around 2kV. For example, the Tektronix TAS220 oscilloscope uses 2kV between cathode and accel­erator grid. Careful design of the vertical deflection plates optimised their curved shape, their length (Ld) and the spacing (d) between them. That, together with a high accuracy wideband solid state vertical amplifier, achieves a working bandwidth of DC to 20MHz. In the next chapter of this series, we April 1996  59 Fig.4: a simplified circuit diagram of an oscilloscope showing the vertical and horizontal deflection amplifiers. will see how post deflection acceleration (PDA) voltages up to 26kV can be used to give a very bright, sharp trace, yet achieve a very low deflec­tion factor of 6.5V/cm and bandwidths up to one gigahertz! A practical oscilloscope Fig.2 shows a simplified high voltage circuit for a small CRO tube, operating at 1.5kV. On a 75mm diameter tube this moder­ate voltage will produce a bright enough trace when Fig.5 (below): a trigger point control circuit. This gives trigger pulse signals at outputs 1 and 2 each time the input signal V(in) passes through some nominated voltage level, V(shift), which you select by potentiometer VR1. 60  Silicon Chip seen in subdued room lighting. The deflection factor is reduced by lower­ ing the acceleration voltage from 2kV to 1.5kV but it is in­creased by using a shorter tube. So we would expect a deflection factor of about 30V/cm. CRO vertical bandwidth is decided by the question: can your vertical amplifier provide enough volts to the deflection plates at the highest frequency you desire? To achieve a screen display 4cm high and 5cm wide, your vertical deflection amplifier must provide a 120V signal swing and the horizontal amplifier must provide a 150V excursion. The author has used a 75mm diameter disposals CRO tube with only 600V acceleration potential, with moderate success. On such a low voltage, the screen trace is less bright or sharp than you desire, yet better than none. A more satisfactory project used a 125mm tube operating on 2.2kV accel­eration, with vertical amplifiers of 5MHz bandwidth – quite useful for TV servicing. In Fig.2 a 60kHz power oscillator excites the primary wind­ing of transformer T1. Secondary winding 1, together with diode D2 and smoothing capacitor C2, generates a 1.5kV DC supply which has its positive end grounded at point F. Its negative end connects through R1 to test point TP3, providing the negative 1.5kV DC supply for the cathode K. The 4V drop across R1 sets the heater slightly more negative than the cathode K, to prevent electron flow from cathode to heater. The resistor string to ground provides a 285V drop across the focus potentiometer VR2. Transformer T2 provides the 6.3 VAC heater supply for the tube. The secondary of T2 is elevated to the neg- ative 1.5kV potential, so it must have at least 2kV insulation rating. Brightness control There are two essential aspects to controlling the bright­ness of the waveforms on the screen. First, the manual brightness control potentiometer VR1 sets the trace to the level to suit the ambient room lighting. Fast rising voltages may need extra brightness to be visible. Second, the timebase sweep circuits must blank out that trace during every retrace (flyback) of the presentation, to prevent confusing patterns. Both these functions are provided by the upper half of Fig.2. Control grid G1 has a 1mm diameter hole through which elec­trons emitted by the cathode may pass. G1 is held more negative than the cathode to control the number of electrons passing through G1 to the screen. Thus, the G1-K bias voltage controls the beam current and thereby sets the trace brightness on screen. In many CRO tubes, a bright (unblanked) trace on screen results when G1 is 10V more negative than the cathode. To block off the electron beam to achieve a dark (blanked) screen, the K-G1 bias must exceed 50V. Secondary winding 2 of transformer T1, together with rectifier D1 and storage capacitor C1, provides an isolated -1.6kV supply (measured between test point TP2 and point A). This nega­tive system finds its ground return via point A, through R2 and a separate +230V supply. For a blanked or dark screen condition, the drive at B to Q1 is made low (around 0V). This cuts off Q1 and causes Q2 to fully conduct, pulling point A down to nearly 0V. That is equival­ent to point A being grounded, so TP2 rests at -1.6kV and test point TP4 at -1.5kV. The brightness control pot. (VR1) has 100V across it. In Fig.2, we set VR1 so that it taps off -1585V, to control grid G1. This potential is 85V more negative than the cathode. With such a large negative bias, the electron beam is completely cut off and the screen is blanked. To unblank the screen, a positive signal of about +5V is applied to point B, making Q1 fully conducting and cutting off Q2. Thus, point A rises to the +75V from zener diode ZD1 and this lifts the complete L2-D1-C1R1-R27 system up by +75V. VR1 still has a 100V drop across it but both ends Fig:6: timing diagram for the trigger point control circuit of Fig.5. are raised by the same amount. Hence TP2 becomes (-1.6kV + 75V) = -1525V; TP4 becomes (-1.5kV + 75V) = -1425V; and G1 becomes (-1585V + 75V) = -1510V. Thus, the G1-K bias is reduced to only -10V, which allows a bright trace on screen. By this means, you set VR1 for the brightness you want on screen. The timebase sweep system then generates a 0-5V control signal at B which automatically blanks out the return (flyback) trace. Note that all these circuits are DC coupled, so that the blanking/unblanking works correctly, even at very slow sweep speeds. At very fast sweep rates, C3 is a speed-up capacitor to overcome delay due to the time constant formed by R27 and stray circuit capacitance to ground. Screen focus To focus a beam of electrons, we pass them through hollow electrostatic fields. This is analogous to the focusing of beams of light by glass lenses. So similar are these two processes that both exhibit the same defects, such as astigmatism and geometri­cal aberrations. In Figs.1 & 2, G2 is the focus grid; sometimes called a focus ring. The small electrostatic field between K/G1 April 1996  61 Fig.7: a rise differentiator based on a 74S00 AND gate package. input signal V(in) passes through the zero axis or at some other point on the cycle. You can adjust the period of the horizontal timebase sweep generator (time/division switch) to display any number (or fraction) of cycles of the input signal. Fig.3a shows about one and a quarter cycles of signal being displayed. The trace is visible on screen from times t1 to t3, from times t11 to t13, and so on. Notice that we do not display every cycle of V(in), because time must be allowed for the beam retrace (flyback) and for holdoff and wait times. In Fig.3, retrace occurs between times t3 to t5 and from t13 to t15. Holdoff Fig.8: this is the timing diagram for the rise differentiator of Fig.7. and G2 acts as a divergent lens. The stronger field (about 1kV) between G2 and G3 brings the electron beam back to a small point on the screen. Thus, you focus the electron beam by adjusting VR2. Potentiometers VR1 and VR2 are elevated to dangerously high voltages and so they are operated by long insulated shafts from their front panel knobs. Astigmatism Astigmatism is the tendency of the beam to come to an elliptical rather than a circular spot on the screen. This is minimised by slightly adjusting the potential on the acceleration grid 62  Silicon Chip G3, by adjusting VR3. That alters the difference between G3 and the average voltage at the deflection plates. G3 rests at about +100V, 1.6kV more positive than the cathode. Triggering To view repetitive signals (ie, a continuous waveform) on the CRO, we superimpose many cycles of the input signal on the screen as shown in Fig.3. To produce a clear display, the hori­ zontal timebase must repeatedly begin its forward sweep across the screen when V(in) passes through the same nominated voltage level each time, as at t1, t11, t21, etc. You may wish the dis­played pattern to commence as the After each retrace is completed, a deliberate holdoff time is incorporated into the system, between times t5 to t6, t15 to 16, etc. The purpose of holdoff is to give the horizontal genera­ tor time to settle and to avoid confused traces when the input signals have a complex period. After the holdoff time, the horizontal timebase waits for the next occurrence of a trigger signal (t11, t21), which ini­ tiates the subsequent forward sweep. The length of holdoff time is dictated by the horizontal generator circuit. It is compara­tively short at slow sweep speeds but relatively long at very high sweep speeds. The duration of wait time is not specified by the circuits; it just depends on how long before V(in) again passes through the trigger voltage level you have selected. Deflection amplifiers Fig.4 is a simplified circuit of an oscilloscope showing the vertical and horizontal deflection amplifiers, trigger point control and rise differentiator. Also shown are the triggered time­base generator and the front panel controls: trigger source selector S2, trigger point control potentiometer VR1, and slope selector switch S1. We’ll start our discussion with the timebase generator which consists of sweep logic circuits controlling a Miller integrator. This generates the rising ramp horizontal deflection signal, by using a selected constant current to charge a low-loss capacitor. The slope of the rising ramp in volts/ second is directly propor­tional to the value of constant current chosen by the time/divi­sion front panel switch, and inversely proportional to the ca­ pacitance value. For very fast sweeps, a small value capacitor is used; larger values of C are switched in for slow sweep speeds. Discharging the capacitor results in the much faster falling retrace (or flyback) signal. The display sequence starts when the trigger signal in Fig.3e triggers the timebase generator. That begins the forward sweep at time t1. Simultaneously, the timebase also generates the blanking signal, Fig.3d, which is fed to point B on Fig.2. At the end of each retrace (t5, t15), the timebase spaces out the holdoff time until t6 (or t16). The system then sits and waits for the next occurrence of a valid trigger signal. The trigger point control unit naturally generates more trigger signals than are used – once each time your input signal V(in) passes through the chosen voltage level. But during forward sweep, retrace and holdoff time, the timebase generator will not respond to those invalid triggers, shown dotted in Fig.3e. Trigger point control Stable triggering of the display is an absolutely essential property of any oscilloscope. To trigger the CRO from your input signal, first set front panel trigger source selector S2 to the INT or Internal position. That will feed amplified input signal from point H to the trigger point control unit. You then set trigger point control potentiometer VR1 to the voltage level at which you want your display to begin. Fig.5 is a circuit which could form the block called trig­ger point control unit in Fig.4. IC1 & IC2 operate on ±15V rails, while Q1 & Q2 work from a single +5V rail for TTL compa­tibility with following circuits. Fig.6 is a timing diagram for Fig.5. In Fig.5, waveform (a) is V(in). Suppose you wish the trace to commence when V(in) passes through voltage level M, on the rising part of the cycle. On the front panel, you adjust potentiometer VR1 to select a DC voltage called V(shift). This is added to V(in) in IC1, an operational adder. Waveform (b) indi­cates the sum of V(in) and V(shift); where we have chosen V(shift) as a negative voltage about half the amplitude of V(in). Thus we call IC1 a level shifter. IC1 is inverting so its output, shown at (c), is just (b) inverted. This signal Because of the small deflection angles achieved by electrostatic means, monoacceleration analog oscilloscope tubes tend to be much longer than their diameter; typically 200-600mm from base to screen. is passed to IC2, an inverting Schmitt trigger. In this condition, IC2 has enormous gain – at least 30,000. So the moment its input, waveform (c), goes the slightest bit negative at time M, IC2’s output saturates to almost the positive rail voltage, about +14V, as shown by waveform (d). IC2 remains in this condition while waveform (c) has any negative value. The moment the input to IC2 (wave- form (c)) becomes posi­tive, at time W, its output switches back to saturation near its negative rail voltage. Any noise on V(in) could make the change over at M and W jittery. To prevent this we add a small amount of positive feedback to IC2. The 100#/10k# voltage divider feeds one hundredth of the output back to the non-inverting input, pin 3. Thus, the moment waveform (c) crosses the zero line, the rise of waveform (d) locks Shown here is a highvoltage DC low-current power supply for the acceleration potential of an oscilloscope. The ferrite core transformer is excited by high frequency drive from a low voltage power oscillator. The high voltage secondary current is rectified and filtered to DC, the large 10kV rated ceramic filter capaci­ tors can be seen at top rear. High frequency primary drive allows the transformer to be light and compact. April 1996  63 Cathode Ray Oscilloscopes – continued If the triggering is switched off, or selected from unrelated sources, the oscilloscope display of a simple sinewave signal can be quite useless, because successive timebase sweeps start with V(in) at different voltage levels. With correct triggering this picture unscrambles to a single trace of six cycles of a sinew­ave. Q2 into saturation, until time W. Q1 is an inverter and level shifter, changing the signal level to a swing between +5V and nearly zero, as at (e). Q2 inverts again to waveform (f). The output of Q1 or Q2 is compat­ible with the following TTL circuits in the rise differentiator. The circuit of Fig.5 is intended only to show the princi­ples of operation. Used with faster integrated circuits, it would work from DC up to moderate frequencies but for a wider passband (eg, 20, 100 or 500MHz) the circuit would be condensed to minimise time delays. Fewer semiconductor junctions, extremely fast tran­sistors and very short leads would be employed. Rise differentiator You have chosen point M on the rising phase of V(in) to be the trigger point. So you want the output of Fig.5, waveform (f), to be changed to a short pulse beginning at time M. Such a pulse can then trigger the timebase generator to begin the forward sweep. But you might change your mind and 64  Silicon Chip decide to trigger the timebase at time W in Fig.6, the same voltage level but on the falling phase. How can the circuits follow your wish? The answer is differentiate waveforms (e) or (f). That can pick off just the +5V rising edge, at time M in (f), or at time W in (e). Fig.7 is a suitable TTL circuit called a rise differentia­tor which actually works by integration, a safe noise-defeating mechanism. This simple circuit uses three sections of a 74S00 quad NAND gate. Its output is a very short pulse coincident with the rising edge of whatever TTL signal is fed to it. Suppose we switch S1 in Fig.5 to output 1, waveform (f) in Fig.6. That signal from the trigger point control unit now feeds ICa in Fig.7 (called waveform L in Fig.8). This is inverted in IC3a to waveform N, which is integrated by R1 and C1, forming waveform P. IC3b then has both waveforms P and L as its inputs. IC3b is a NAND gate, so it gives a low output only when both its inputs are high. But observe in the timing diagram that, due to the R1C1 time constant, P does not drop immediately when waveform N does, at time M. Rather, P takes a small time after time M to fall from its +4V output. So depending on the values of R1 & C1, P is still above the TTL threshold level (+2V) for a brief period Delta(t) after time M. During that very short interval, Delta(t), P and L are simul­ taneously high (in TTL terms). That is, (P.AND.L) is a logical high signal for that brief time, as the timing diagram shows. IC3b promptly inverts this to a logical low (waveform U). IC3c inverts again, giving waveform Z, a signal at TTL high level for a short period from time M to M + Delta(t). This is wave­ form (g) in Fig.6, a pulse suitable for triggering the timebase generator. Suppose now you change your mind and wish to trigger the oscilloscope at that same voltage level of V(in) but on the falling phase, as at W in Fig.6. In this case, you just switch S1 in Fig.5 down to output 2, selecting waveform (e) in Fig.6. This now becomes input signal L to IC3a in the rise differentiator, which detects the rise of waveform (e) at time W. As a result, it gives forth its trigger pulse every time V(in) passes through the chosen voltage level but on the falling phase. Most oscilloscopes provide a wealth of trigger sources such as External, 50Hz Line, Single Sweep and Auto, triggered by an internal free-running flipflop, so there is always some display on screen, with or without vertical input. Others commonly found include TV Horizontal, TV Vertical, DC/AC Coupling and Noise Rejection. Next month we will look at post deflection acceleration (PDA), calibrated screens, deflection amplifiers, probes, time­base generators, shift controls and dual timebases. Acknowledgements Thanks to Philips Scientific & Industrial and to Tektronix Australia for data and illustrations; also to Professor David Curtis, Ian Hartshorn, Ian Marx and Dennis Cobley. References “ABC’s of Oscilloscopes”; Philips/ Fluke USA. “Solid State Physical Electronics”; Van der Ziel, Prentice Hall NJ. “XYZ’s of Oscilloscopes” and AppliSC cation Notes; Tektronix Aust.